CA1228436A - Code division multiplexer using direct sequence spread spectrum signal processing - Google Patents

Abstract

CODE DIVISION MULTIPLEXER USING DIRECT
SEQUENCE SPREAD SPECTRUM SIGNAL PROCESSING

Abstract A plurality of transmitters synchronized to a common clock each transmit a data signal spread by a common bipolar pseudo-random code having a different predetermined assigned code sequence clock. A receiver, synchronized to the clock, discriminates the signal transmitted by a predetermined transmitter from signals transmitted by the others by generating a first bipolar pseudo-random code that is a replica of the common bipolar pseudo-random code and has a code sequence shift corresponding to that of the predetermined transmitter, and a second bipolar pseudo-random code and has an unassigned code sequence shift. The difference between the first and second bipolar pseudo-random code sequences, which is a trinary code code sequence, is cross-correlated with the incoming signals. The cross-correlated despreads only the signal applied by the sequence having the predetermined code sequence shift. Each receiver includes a number of correlation detectors offset from each other by a fraction of a code chip together with decision circuitry to identify cross-correlated peaks for optimum synchronization.

Description

~.347/391 Canada CODE Division MULTIPLEXER USING DIRECT
SEQUENKS SPREAD SPECTRUM SOGGILY PROCESSING

Technical Field Ike invention relates generally to code division multiply my using direct sequence spread spectrum signal processing, an more particularly, toward signal processing to increase the number of ~ransnitters multiplexed for a given code length.

auctioned All In a spread spectrum Sistine, a transmitted signal is spurred over a frequency band that it much wider Han the minimum bandwidth required to transmit particular information. -whereas in other forms of modulation, such as amplitude modulation or frequency modulation, the transmission bandwidth is comparable to the bandwidth of the information itself, a spread sFectr~m system spreads an information bunked of, fox ex3mp1e, only a eye kilohertz over a Rand that is many megahertz wide, by modulating the information with a sideband encoding signal. Thus, an important characteristic distinguishing spread spectrum systems from other types of broad band transmission system is that in spread spectrum signal processing, a signal other than the information buying sent spreads the transited signal.
Spreading of the transmitted signal in typical spread sperm systems is provided by (1) direct sequence modulation, I frequency hyping or (3) pulsed-FM or "chirp" ~cdulationO In direct sequence modulation, a carrier is modulated by a digital code sequin ox whose bit rate is much higher than the information signal bandwidth. Frequency hopping involves shifting the carrier frequency in discrete inurements in a pattern dictated by a code sPquenc~, and in chirp modulation, the carrier is en o'er a wide band during a give pulse interval. Other, less connately used, carrier spreading techniques include time hosing, wherein ~ransmissicn time, use of a few duty cycle and short duration, is governed by a code sequence and tome frequency hoping wherein a code sequence determines both the transmitted frown and the time of ts3nsmission.
Applications of spread spectrum systems are various, depending upon characteristics of the cedes being elude for band spreading and other factors. In direct sequence spurred sickroom systems, for employ, where the code is a pseudo-rardom Swiss, the composite signal acquires the characteristics of Lucy, making the transmission undiscernible to an eavesdropper who is rot capable of decoding the transmission. .~dition.~l ar~lications include navigation and ranging with a resolution depending upon the particular cede rates and sequence lengths used. Reference is made to the textkcok of ARC inn, Spread Sickroom Svste~s, John Wiley and Sons, New York, 1976, especially __ chapter 9, for application details.

LO I

Direct sequence adulation involves modulation of a carrier by a code sequence of any one of several different formats, suck as AM or FM, although buffs phase-shift keying is the most common. In buffs phase~shlft keying (SKYE, a balanced mixer s whose inputs are a cede sequence and an OF carrier controls toe carrier to be transmitted with a fist Phase shirt of X when the code sequence is a I and with a second phase shift of (180 + X) when the code sucrose is a Jon. Buffs phase shift keyed modulation is advantageous over other forts because the carrier is su~Dressed in the transmission waking the transmission more difficult to receive by conventional equipment and preserving more power to bye applied to information, as opposed Jo the carrier, in the transmission. Cbarac~eristics of buffs phase-shift keying are given m Chapter 4 of the aforemen~iored Dixon text.
the type ox code used or spreading the bandwidth of the transmission is referable a linear code, particularly if message security is not required, and is a maximal code for best cross correlation ~aracteristics. .~3ximal codes are, by definition, the longest cedes that could ye generated by a given shift register or other delay element of a given length. In binary Choctaw register sequence generators, the maximum length (ML) sequence that is capable of bPinq generated ox a shift register having n stages is on _ 1 bit. A shift register sequence generator is furrowed from a shift register with certain of the shift register stages fed back to owner stages. The output bit stream has a length depending upon the number I stages or the register end feedback erwloyed, before the sequence remelts.
shift register having five ages, for example, is capable of generating a 31 bit binary sequence (i.-. 2; - 1), as its maximal length (My sequence. Shill register Lo sequence generators having a large number of stages generate AL sequences that repeat 50 infrequently that the sequences appear to be random, acquiring the attributes of noise, and are defoliate I

detect. Direct e~qU~C2 systems en_ thus sometimes culled Sudanese systems.
properties of maxim sequences ye summarized Lo Section 3.1 of Dixon art feedback co M cations for maximal code generators from 3 to lC0 stages are listed in Table 3.6 of the Dixon text. For a 1023 bit code, corres~ondLng to a shift resister having 10 stages with maximal length feedback, these are 512 lo and 511 "owns; the difiarenc~ it L whereas the relative positions of lo and tons very among ML code sequences, the number of lo and the namer of "Ohs Lo each maximal length sequence are constant for identical ML
length sequences.
Because the difference between the number of lo end the number or Us Lo any maximal length sequence is unity, auto correlation of a maximal linear code, which is a bit Joy bit I cc~arison of the sequence with a phase shifted replica of itself, has a v lug of -1, except at the 0 1 bit phase shift area, in which correlation varies linearly Exam -1 to (on - 1). A 1023 bit Maxwell code (on - 1) therefore has a peak-to-average au~ocorrelation value of 10~4, a range of 30.1 dub.
It is this characteristic which makes direst sequence spread spec*2um transmission useful in code division multi~lex~ng.
Receivers set to different shifts of a common ML code are synchronized only to transmitters ha m g that shift of the common code. Thus, more than Gone signal can ye unambiguously transmitted at the tame frequency and at the same time. In an autocorrel~tion toe multiplexed system, there is a cclmon clock or liming source to wow several transmitters and at least one receiver are synchronized. me transmitters generate a common maximal length sequence with the code of each transmitter phase snifter by at least on bit relative to the other codes. The receiver generates a local replica of the gammon transmitted axial length aequenC
having a ccae sequence shift that corresponds to toe swift of eye particular transmitter to winch the receiver is tuned. Toe locally generated eons is autoc~rr~lat~d with toe income my I

signal by a correlation detector adjusted 50 as to recognize the level associated with only ' l-bit s~mchronization to desp~ead and extract information prom only the soggily generated by the predetermined transmitter.
Because the auto correlation characteristic of a maximal length code equines has an offset corresponding to the inverse of the cede length, or Van I) where V is the agony or voltage corresponding to I and n is I the number of shirt register stages, overlap occurs between neighboring channels. Thus, there is imperfect rejection of unwanted incoming signals. Unambiguous signal discrimination thus requires a guard band between channels, reducing the number of potential transmitters for a given code length. A long maximal length sequin ox compensates for the guild band to increase the number of potential transmitters, but this wow synchronization and creates power imb~lancs of the multiplexing transmitters.

Disclosure of I've lion It is, accordi~gl~, a primary object of the invention to provide an in,Droved auto correlation type code division multiplex method and system wherein an increased number of transmitters log a given code length can be unambiguously disc.~minated. Another object is to provide a code division multiplexing method and I system using relatively short AL codes to discriminate among a relatively large number of transmitters.
These and other objects are satisfied my the method and system of the present invention, wherein a plurality of transmitters and at let one receiver are synchronized to a common timing signal source. Each transnit~er transmits a data modulated carrier spread by a by w far pse~do-randcm code Nash is a different assigned shift of a kimono bipolar code sequence. To discriminate the signal transmitted by a ~r~aetermined transmitter f an the signal transited by the others, the receiver Abner toes 8~3~

two focal bipolar ps~udo-random codes that are replicas of the transmitted common bipolar pseudo-randcm cede. One of the locally venerated codes has the same code sequence shift as the code sequence shift assigned to the predetermined transmitter; the other locally generated code has a code sequence ski t that is not assigned to any of the transmitters. Ike two locally generated codes are processed to obtain a urinary code sequence which is cross-correlat~d with thy ir.ccminq signal to discriminate and extract information from the code transmitted by the prede~rmined transmitter.
In accordance with another aspect of the invention, the binary code sequence may be a maximal length (AL) sequence having good auto correlation properties. Information may be embedded within the binary code sequence using phase shift keyed PER
swigger modulation.
Still other objects and advantages of the present invention will become readily apparent to those skilled in this art from the following detailed description, wherein there is shown and described only the preread embodiments of the invention, simply by way or illustration of the best modes contemplated of carrying out cur invention. As will ye realized, the invention is capable of other an different e~tod~ments, and its several details ace capable of modification in various, obvious respects, all without departing prom the invention. Accordingly, the drawings and description are to be regarded as illustrative in nature, and net as restrictive.

Lo Lo , grief Descrl~tl of the Drawings Figure 1 is a simplified block diagram showing a DOSS I e division multiplex receiver;
Figure 2 is a representation of a bipolar ~seudo-random pulse sequence;
Figure 3 is a diagram showing an cut x correlation pattern for a bipolar pseudo-random pulse sequence or the type shown in Figure

2;
Figure 4 is a superposition of several auto correlation patterns corresponding to neighboring transmitters in a code division multiplex siesta;
Fissure 5 is a diagram corresponding to Figure 4, with signals of neighboring transmitters separated by guard bands;
Figures Audi are wave forms showing urinary code lo generation;
Figure 7 is a simplified block diagram skewing a receiver operated in accordance with the principles of the invention;
Figure 8 is a diagr3n skewing an ire lived cros~-correlation pattern button a locally deviled urinary cede sequence and an incoming binary code sequence in accordance wit the invention;
Figures awoke are diagrams skewing correlation patterns developed by multiple Connally correlation detectors in accordance with various embodiments ox the invention;
Figure 10 illustrates an actual correlation pattern obtained in the receiver of the present invention when operated in the presence of various degrading factors;
Figure 11 illustrates an analog enbcdiment of multiple correlation detectors or determining the degree of correlation in accordance with the invention Figure 12 is a circuit amplification of the analog embodiment of ire 11 using binary reference signals;
Figure 13 is a further circuit simplification of the analog circuit ox Figure 11, using digit logic to reduce toe number of analog multiplexes;

Figures aye and 14(b) illustrate Jo methods or implementing the circuit of Figure 13;
figure l; is a digital implementation of one channel of the circuit shown in Figure 11;
Figure 16 is an N-channel qeneraliza~ion or the circuit implementation in Figure 15;
figure 17 skews another digital implementation of a single channel correlator of a type shown in Figure Lo;
Figure 18 is an Connally generalization of the circuit shown in Figure 17;
Figure 19 illustrates an Ln-phase end quadrature-p~ase correlation pattern, together with the locutions of sub-r~c~iver channels for correlation detection;
Figure aye and 20(b) are flow charts showing two Lo alternative ~cthcds for performing fine tuning of the receiver;
Figure 21 illiterates a microprocessor eased circuit for performing fine tuning of one receiver and signal presence detection;
Figures aye and 22(b) are flow charts respectively skewing methods for correcting receiver t and for purify signal presence d t~cti~n;
Figure 23 is a flaw chart showing one technique for performing coarse tuning of the receiver;
Figure aye) - eye are tiring diagrams stowing the I relationship of lining pulses between a transmitter and a receiver;
Figure 25 illustrates a circuit fox locking a transmitter and receiver to the save timing pulses; and Figure 26 illustrates a microprocessor eased circuit or Perform m g data recovery in the receiver.
B _ General In sued spectrum communications, spreading of 5-gn~1 bændwidtn beyond the bandwidth normally required for data being I

transmitted it accomplished by first phase shift keyed (PER) modulating a carrier waveform by data to ye transmitted, and then modulate my the resultant signal by a reference pseudo-randcm code of length L running at a repetition rate which it normally at least twice the data rate. Forms of modulation other than PER can be applied to modulate the carrier as s to spread the composite sign 1, although PER is preferred for no owns sex forth earlier.
To d2mcdulate the ion 1 transmission, the r æ wived signal it heterodyned or multiplied by the some reference code as the one used to sired the composite transmission, and acumen that the transmitted and locally generated receiver codes are synchronous, the carrier inversions caused by the cede PER modulation at the transmitter are removed and the original base-band modulated I carrier is required in the receiver.
Figure 1 illustrates the fundamental elements of a basic s~re31 spectrum receiver incorporating one aspect ox the invention. Receiver 100 receives a direct sequence spread Spectrum (DOSS) signal transmitted by a particular transmitter Mooney a plurality of such t~ans~itt~rs, and p~qcesses thy received signal to discriminate the signal transmitted by the particular transmitter from among the signals transmitted by all the ~ransnitters. Lear my in m m d that the received signal is essentially modulated twice, that is, the carrier is modulated with data and then the ccmposit~ is modulated by a pseudo-random code sequence to spread the composite over a bandwidth that is comparable to the bandwidth of the pseudo-random skunk, rec_lvex 100 provides two stages ox demodulation of the received signal to extract the transmission data. The received DOSS signet is first het~rodyned or multiplied by the c ye of the particular transmitter whose signal is being discriminated from among the others. Thus, assuming that the codes generated at the transmitter and receiver are synchronous, toe carrier inversions caused by the code So modulation at the transmitter are roved I

-- Jo -at ~ltipl~r 102, and toe original b band modulated courier is restored. The narrow-barld restored tarrier to I d to a band pass filter snot shown designed to past only eye base-band ~ulatQd carrier. 3ese-~and date are then extra by hotter or multiplying the restored carries by a locally generated carrier at ~ltip.lier I Thy ought of ~ltiplier 104 I alkali to a convoy tonal correlation 106, such as awl ;ntegr~b~ and dump circuit, followed by a ample end hold circuit idea develops signals cDrres~on go to the it'd data.
the receive_ 100 is controlled key a sty rod micro,~rccessor last s~nc.~ronized to a system clock lo, to Nash ye transnitt~rs are also synchronized. ~ecauxie noise and undesired t~ans~igsio s are treated in the same process of multiplication in multiplier 102 my the Luke generated reference cede Tut c~m~resse_ toe I wived direct sec3~enca signal into the original ~Lrri r width arty ins signal not ;,yn~onou~ with the locally junta Rufus code is Seward into a 'aridity equal to the sun of the bandwidth ox the go sign!. and the bandwidth of the r_ferenca rode. Since this synchronized L'lpUt signal it zoo ma into a bandwidth tilt is a Lucy a wide a the rs~er2nc~
code, a band pass fulcra can rocket a signi~ice~t I of toe per of an desired soggily. this is ye 3ignificas~.c_ ox a SASS
siesta ch~oncus is~?ut signals at the refQr&ncs coma ~:dslat~d bandwidth are transfo~ed to the 'oase-band adulated b3n~widt~, whereas run ours iota ~ign~l~ rain Syria over thy ccde-r~dulated bound.
Synchronization ~rccessins maces use or a property inherent in ye attic cod at is elude at eye trueness he autocorr~latlon of a sisal length (~) swoons, ~ltiplicaticn of toe sequence by a to sniff rollick of itself, is at a hen svnG~onization i achieved and has on absolute v UP await dross JO I none it ye ;~2gnir!de of the corn sequence and L it ye cue lo s svr.cr~o~ lien buckeyes lost it -ye Lowe dif-~r~nco Byron ye cock end lo replica approaches a code chip or greater). me sign of the autccorrelation pattern is dependent upon the data bit being used to modulate the transmitter. It is thus possible to recover the transmitted data at the receiver by monitor my the sign of the autccorrel3tion output when the receiver and transmitter are properly synchronized.
Referring to Figure 2, a pseudo condom code sequence of a type to which receiver 100 is tuned is bipolar, that is, it is zs~umed to switch polarities of a constant voltage power supply.
In the invention, bipolar, rather than unpiler, s~uencas are used to improve power transmission efficiency, since the carrier is suppressed in bipol r transmission. Bipolar transmission also avoids high conc~tra~ions of energy in any frequency band to help avoid interference between transmissions by different transmitters in the system. Each bipolar sequence has a magnitude P and a chip duration To. The length of the ML sequence defends upon the number of Defoe en transmitters whose signals æ e to be code-division mNlti~lex~d within the system Each transmitter is assigned the same transmission code having a different specified chip ox the cc~mon AL sequence. The maximum number of transmitters that are capable of being multiplexes within this system thus corresponds to the length of thy AL sequ~nco.
the number of transmitter that may be multiplexed without interferon ox within a code-division multiplex siesta or this type is equal, theore;ic~l~y~ to toe bit length ox the sequence. For an LO code having a length of 63 bits, for example, the transmission channel it theoretically capable of multiplexing 63 differ nut transmitters. This assures that svncnronization is deemed to be achieved between the receiver and a preselected transmitter when the autocorrela~ion between the code received or the transmitter and the focally generated code, both synchronized to a cc~mon tiring uric, is at a eke In practice, however, the number or t ans~itt~rs that can go code division multiply in the system is much lower than toe it theoretical maximum, 'c2cause there is overlap between neighboring correlation curves due to the -P AL term m the auto correlation of the AL sequence. This can ye wetter appreciated with reference to Figure 3 which shows a correlation curve for a single transmission and Figure 4 which skews a number of correlation curves for n~ighboe~r.g transmissions, that is, or transmissions that are time offset from each other by a single code chip.
In Figure I, the correlation curve has a magnitude -PLY
when the transmitted and locally generated code sequences no time off from each owe by greater than a code chip To, where P
is the absolute magnitude of the quince and L is the sequence length in bits. When toe tsa~smitted and locally generated codes are noes synchronization, that is, are within a lime offset of one code clip of each other, the correlation increases Lo magnitude to l; a weak of pi at perfect synchronization. Thus, synchronization between the receiver and a single transmitter can lo detected by monitoring the correlation output and deeming synchronization to exist wren the correlation signal is above a predeternin~d positive value.
I Referring now, hcw~vert to Figure I assume that there are three transmitted cede sequences k, k-l and clue, tome shifted from each other by a single code chip Each correlation has a positive peak value of and a negative peak value of -32/L, as in Figure 3. the correlation curves ox neighboring code sequences overlap, within the regions shown by cross hatching in Figure I.
In those resins, neighboring cede sequences have gammon correlations, making it im~cs~ible to distinguish Boone transmissions. As a practical tatter, to avoid interference between t~ansmisslons, it is necessary to insert a guard band between sequences, as shown in Figure 5. mix is provided by assigning transmissions to sequence shifts Coors ending only to alternate cede chit allays, rather than to ever code chit delay as in issuer 4. The result is that, at cost, only one-h~lf the number an tsa~smiasions, ccm~ared to the theoretical maximum number, can be multiplexed. In tactics even fewer than one-half the theose~ical maximum transmitters eye capable of bins ~wlti~lexed in a cede division ~wltiplex system swig bipolar sequences because a guard bond thaw is greater Han that provided S using only alternate code stilt delays is required to avoid synchronization ambiguities.
}n accordance with one aspect of tie invention, the number of transmitters that are capable or being multiplexed is increased Jo one less than eye theoretical limit by cross-correlating the input signal with a urinary code developed by obtaining the difference between the co e seqyencs assigned to vie particular transmitter Jo which the receiver is tuned and a code sequence what is unas3ignedo In other words, Jo bipolar code sequences no developed at the receiver. One of the codes is the replica of the I gammon code sequence transmitted by all the transmitters and has a sequence shift Nat corresponds to tune sequence shift ox a redetermined one of the transmitters. The second code is a replica of the common bipolar sequence and ha a code sequence shirt that is not assigned to any ox the transmitters. One of the locally generated codes is subtracted prom the other, and the resultant, which is a urinary code sequence, is correlated Wit the incoming signals. the sequence shift of the urinary code sequence is brought to Within one code chip of the sequence generated by the preselect æ transmitter, using a static s~mchronization technique to ye described below. perfect synchronization between the receiver and preselected tan sitter is obtained using dynamic synchronization, also to ox described in detail below, obtained generally by successively shifting the timing of the receiver ox a fraction ox a code chip and monitoring the output of the correlator. When the correlation output is at a peak, the receiver and preselected transmitter are considered to be synchronized to each other. Assuming new that the receiver and transmitter are also synchronized to corresponding clock pulses (i.e., the transmitter is not synchronized to one clock pulse arid the receiver synchronized to another), the polarity of the correlation output is monitored to extract Tao transmitted data.

~Z~Z8~3~

Development or the truer pulse sequence to be cross-correlated with the transmitted sequences is better understood with referent ox to Figures Audi In Figure I, a transmitted bipolar sequence so having an absolute magnitude P
and chit period To is shown. This sequence is a simplific lion of an act sequin ox which, in practice, would be substantial longer, e.g., 53 bits. Within the reoccur is developed a first reference pulse sequence f shown in Figure I. Ike sequence f is identical by the seouencP sty transmitted by toe redetermined transmitter shown in Figure I, because the transmitter and receiver sequences have the same delay and are presumed synchronized to each other.
The receiver generates a second reference pulse sequence c, shown in Figure I, winch is the same sequence as the one transmitted by the preselected transmitter as will as by all eye other transmitters but has a sequence delay what is not assigned to any ox the transmitters.
my difference k (t) - of between the Jo Luke generated reference pulse sequences is ~btaLned, to Roy the trainer plus skins shown in Figure I. Toe urinary sequence has a value [+2, 0, I depending upon the relative bin y values of the two reference pulse sequences f and c.
It is to be understood that the sequence length in the example shown in Figure 6 is bits, although Lo practice, much 25 , longer sequences would be applied to acccm~odate a relatively large number of transmitters to be code division multiplexed Referring to Figure 7, de~Jelo~ment of the urinary reference sicken to be cross correlated wit incoming bipolar pulse sequences for signal demultiplexing is roved in a receiver 200. The receiver 200 receives the transmitted Pulse sequences so and applies the inc3ning sequences to the inputs of a first correlation multiplier Z02 and a second correlation multi~ller 204. The first correlation multiplier 202 multiplies the incoming sequences so ox the locally g~lerated reference pulse errs ~1.2~3~

f having a equine shift corresponding to toe sequence swift of the preselected transmitter. The multiplier 204 multiplies the oncoming Skunk so by the pulse sequence c having an unassigned pulse sequence shift. Thy resultant multiplication S products are applied lo a difference circuit 206, and the difference is integrated and sampled in a standard correlation jilter 208 to develop on output signal Youth It is pointed out that in Figure it the input sequences so are first multiplied respectively by the two reference pulse sequences f and c, and then the product difference is obtained in difference circuit 206. This is equivalent to obtain no the difference between the two reference pulse sequences f and c and then multiplying the d furriness by the incoming sequences so I The resultant cross-correlation is shown in ~igurs 8. Not that each correlation curve has a value 0 when the preselected transmission and l~c~l1y generated reference sequence rye are displaced prom each other by more than one cede chide This contrasts to the cross-correlation curie of Figure 3, wherein pa there is a negative residual correlation having a magnitude pi m e magnitude of the correlation curve increases linearly to a peak value of PLY L when the preselected transmitted and lcc?11y generated reference pulse sequences are synchronized.
the advantage of this correlation strategy is a recrated 'Dye comparing Figure pa skewing the correlations of a number of neighboring transmissions in accordance with the invention and Figure 4. In particular, Fig. pa shows codes with a Syrian of 2 code chips. Dover, it will be a~oreciated what the figure pa transmissions can lo displaced from each other my a single code shift and that there is no overlap Boone the correlations of adjacent transmission, whereas in Figure 4, overlap ox us in the cross-hatched portions. The invention thus enables the number of transmissions c3Dable or icing multiele~ed to be equal to one 1QSS
than the length of the purse Sirius in bits, result await is 134~

no possible using prior art systems. Even it a guard band is placed ennui transmissions in the strategy shown in Figure pa, the number ox transmissions that can be reliably multiplexed is substantially great-r than the number that can be reliably multiplexed using the correlation strategy shown m Figure 4, Assure that the coda-division multiplexed PER signal v incoming at the receiver is expressed as follows:
v PjdiXj(t)cos(W5t O) + I (1) where for J incoming transmissions:
O t T, where T is a cede chic Jo nod;
Pi is the power within each incoming bipolar pulse sequence;
do us the polarity or sign ox each corresponding incoming sequence;
Xj(t) is the transmitted data;
We is the frequency of the carrier in radians;
O it the carrier phase; and No is noisy.
The output VIA of the conventional receiver, using 3 single reference code sequence, is defined by the oiling:
1 AL, VET Prudery L ~JPjdj + (2) jar where:
Pry is the power of the desired incoming sequence dry is the data sign of the desired sequence;
L is the pulse sequence length Lo bits;
Pi is the Corey of each of the undesired sequence;
d. is the corn sponging data sign of the undesired sequence; and 30 is Boise.
The output VET of the receiver operating in accordance wit the principles of the invention is defined as follows:
Vet Pry; (1 + l/L) (3) because the correlation method of the invention involves a subtraction ox a code sequence having an unassigned code equines shift, ill undesired transmission components (iaantified by the subscript no in the output VET are perfectly rejected, whereas on the prior art receive, the output VOTE involves contributions of the undesired transmissions (having the subscript ) as well as the desired transmissions (subscript "r").
Multiplexer urinary signal correlation inducts an additional Thor decibels of degradation on data signal-~o-noise demodulation with respect to white noise fearing at thy receive input cam eased to conventional correlation using only the particular transmission binary pulse sequence Thus, No = O
(~) 'I
2(1~1/L) .

m e multiplexing strategy discussed above results in perfect unwanted access rejection capability using ML cocos or any length in a c~de-division multiplex system. In the past, only AL codes of sufficiently long length were potentially usable with the number of allowable m1~1tiplexe~s being much less than the code length. Even therm, power ~mkalances of the multiplexing transmitters occurred.
Additionally, the ideal cross correlation pattern in Fig. pa lends itself to multiplexing s homes using Gore than the theoretical limit of code, each time-offset by less than a code chit, and assuming a Gore complex receiver confisurationO For example, it has been discovered that the number of transmitters which could be multiplexed can ye increased to 2 x (L-2) channels by adding a code between each of the code sequences shown in icky.
4, with only a slight trade off in overall receiver signal-to-noise performance. so Jowl in Fig. 9b an additional code can ye inserted eighteen each of the cedes shown in Fig. I.
The codes are detected at a ol~rality of teds roved at the receiver. Toe outputs of the various river taps shown in jig.
9b are us follows:

~22.~3~36 ABLE I
1. extra code 2. 1/2 extra code

3. null

4. l/2 code 1

5. code 1 l/2 code l'

6. 1/2 code 1 + code 1' l/2 code 2

7. 1/2 code l' cede 2 l/2 cede 2' The sequence of equation. yo-yo then lo solved log each channel:
channel 1 = 2 x zap 4 channel 1' - 2 x (zap 5 - charnel 1) channel 2 - 2 x Tao 7 - channel 1' - Tao 4) channel 2' Jo 2 x trap 7 - channel 2 - tap ;) I n n I

channel L' - 2 tupelo 3) In practice, the avow arrangement would be squint difficult to implement due to Roth noise and synchronization problem... An alternative implementation Gould require that a no of the carriers occurred at the onto where the correlation envelope is equal to 1/2 the maximum In such an arrang.~men~, the equations for the Cutouts of the taps become:

LOWE II
1. extra code 2. null 3. null 4. null 5. code l 6. code l' 7. c ye 2 This arx3ngement allows or full data Riviera without interf2r~nc~. however, it may swill ye ought susceptibly Jo noise.

8~3~

In order to overcome the above orobl~s, there is shown in Fig. 9c an arrangement in which two or Gore code sequences are grouped together and separated by guard-bands. the exact succession of the groups or the patterns co~prisiny the groups is independent of this arransem~Lnt. This approach also allows the grouping of transmitters with similar characteristics and simDli~ies synchronization problems.
Any additional modulation by data bearing signals and that necessary for ID roved communication batten ~ransmltter~ and receivers can be incorporated in the above described strategies.
The only condition required is that any additional modulation must not destroy the necessary timing ox the skirted pulse sequences thereby mum tenon receiver multiplexing sensitivity.

I Svnchrcnization - General m e rev river and preselected transmitter must ye time synchronized to each other before data can be extract Sweeney that the receiver and transmitter are synchronized to a common Tim my source (if the commerce power line it the transmission medium, ccrmon Managua can ye obtained prom the I Hertz power source, synchronization is a tatter of dotting receiver Tim m g to different proration delays ox the transmitted signal as well as to the timing signal and to delays inherent in the transmitter and receiver Same of these delays are fixed, and can ye compensated using a "static" delay, to synchronize the receiver and predetermined transmitter to within one cove chip of each other, wherein a chip is defined as thy bit period or the pseudo-random code generator.
In general static delay can be compensated during initial calibration of the receiver, since sty static delays ens fixed.
difficulty occurs, however, wren the transmission medium is a transmission line with the transmitter end receiver synchronized to a cordon timing Coors, and wherein communication ennui the owe units is bidirectional. Static delay just Tess be amend ~Z~3~36 from Jo reference points, one where the transmitter is at the timing source and the other where the receiver is at the timl~g source.
With the transmitter located at the timing source and the receiver located elsewhere, the Tim my signal and transmitted signal will propagate at aDproximat~ly the same steed from he transmitter to the receiver. Other timing variations between the transmitter and receiver are due to delays induced within the transmitter and receiver circuitry, and con be preset to synchronize the transmitter and receiver to within one code chip of each other. ~11 receivers remote rum the Managua source can thus have identical static delay .
If the receiver is located at the timing source and the transmitter is located elsewhere, however, each receiver Jay l; require a static delay that is unique for each remote transmitter to account for different signal propagation distance. Thus, to enable a receiver to receive signals from a multiplicity of transmitters, the static delay of the receiver must key variable.
In practice, the static delay between each transmitter and the receiver is measured upon installation of the transmitter; what static delay value for all future communications with a particular transmitter is onset within the receiver. Ah never a transmission is received from that transmitter, to obtain united syncbroni~ation of the transmitter, receiver timing is automatically adjusted Jo accommodate the delay associated with the particular transmitter.
In one embodiment of the invention, there are a plurality of transmittes/receiver units disposed in a sucked "~aster/sLave'' arrangement. In this arrangement, one transmi~ter/receiver unit, aloud the master station, acts as the source of timing signals for the other stations (slave units). The amount of delay associated with the t mint signals Byron the master station and each of the slave stations includes such things as ye filter delay for the timing signal source ye the master station, the I

received filter delay at the master station, the signal propagation delay between the ester and a particular slave, the coupling delay a the master station and the transmit flutter delay at the master station. Rno~led~e of those various delays will give an estimate ox the amount of stat c delay associated Boone the master station and a particular slave station. however, some variation in each delay will occur wit changes in the transmission line associated with temperature changes, transmission frequency, etc.
While dynamic delay adjustments can take care ox most of these changes in the static delay characteristics between the master and slave units, the multiplexing capabilities or the system may be somewhat reduced because the receiver at a ~articlllar master or slave unit must be capable of track my delay variations over a range of several code chips. This requires a guard band that is wide enough to allow the signals of two adjacent receiver to vary in time over their associated bands without interference.
cover, it has been discovered that the amount of required guard band may be reduced by periodically measuring, at the master station, the static delays associated with signal transmission between the master station and each of the slave stations and then Periodically adjusting the transmitter signal timing at the slave in order to bring tune static delay back into a desired range.
This allow more slave stations to transni~ at one time since one guard band requited for delay variations can ye greatly reduced thus allowing Gore usable code delays 'or multiplexing variations from synchronization established by the static delay are compensated ox a dynamic delay mechanism within each receiver. The dynamic delay consists of two stages: fine tuning and coarse tuning. whereas static delay timing causes the receiver and predetermined transmitter to ye synchronized to each other to within one code chip fine tuning uses correlation detection to make fine adjustments in receiver timing as function of received transmission, rather than as a function of an expected transmission (static delay).
After fine tuning has established that receiver timing is at a local correlation Fake it becomes necessary to determine if the S local peak to which the receiver is timed is the noorrectn local peak for best correlation This is necessary because, depending upon the correlation properties of the code selected, well as other factors, there are likely to be multiple correlation pucks, with the primacy local peaks hove the greatest peak magnitude.
These multiple peats arise Roy carrier correlation within the ~lTC code correlation peak. Finally, it must be determined which of the Sistine Tim my pulses present in each day bit is the prover one for synchronization. Without such a determination a condition can exist where m the ~ansmitter is tacked to aye lining pulse while the receiver is Lockwood to another timing pulse. This is because these are two timing pulses m a data period and incorrect timing causes a quadrature condition between ~ranS~ittQr and receiver data periods. Thus, the net energy for such quadrature data periods is zero, Even with the receiver and transmitter properly synchronized to each other, data cannot ye extracted from the received sequence because it is not possible to detect and decode the data transmission unless the receiver and transmitter are locked to the same timing pulses Fine tuning and coarse tuning as well as synchronization to the roper tiring pulse within each data bit hall now be described in more detail.
Figure 10 illustrates the correlation pattern obtained by cross-correlati~g an Lncaming~ bipolar Sue sequence together with its carrier and the luckily generated urinary reference sequence. the correlation pattern nay a major peak at receiver timurg V1 and has minor correlation peaks at receiver timings V2, TV V6 and V7, referred to hereinafter as ''channels" The correlation peak at primary channel Al depends upon the correlation properties of the code selected as a finch on of code clip time delay difference between the inccmir~ code sequence end ~2~36 the reference code sequence. the correlation is at a peak when synchronization between the receiver and transmitter is achieved, with the absolute value of the correlation dsoFping to zero as the synchronization difference approaches a code chip or greater. It should be noted that, due to imperfect correlation refreshes of the cede and due to the influence on correlation by the sinusoidal carrier, the correlation shown in Figure 10 is approximately sinusoidal as compared to the piece-wise linear, ideal correlation profile shown in Figure pa which does no include a carrier. This is the reason that coarse tuning is required; fine tuning adjusts receiver timing until a correlation peak is determined, coerce tuning then determines whether the correlation peak is the major correlation peak associated with channel Al or is a inure correlation Flak associated with channels I V3, TV or V7, ox 1; others.
In crudeness with one aspect ox the invention, synchronization of the receiver is achieved by providing a plurality of seDarato sub-rece~ers or correlation detectors that are tuned to each receiver channel. Assuming that each of the channels Al, V2, V3, V6 and V7 aye spaced apart from each other in time by one third ox a cede chip, wine tuning adjusts the receiver timing such that the channels are ail located at local peaks.
Furthermore, assuming that channel I is ~ichin a code chip of being synchronize, the channel Al is within one sixth of a code chip of a focal pea. The outputs of the correlation detectors are applied to a microprocessor 314, described below, to develop a receiver t~mir.g signal for synchronization to the transmitter and to extract transmission data. Various embodiments of the mull-ipl~
correlation detectors are illustrated in Figures 11 Correlation Detection One embodiment of the multiple channel correlation detector shown in Figure if is generalized for correlation Guerrilla. me multiple channel correlation circuit identified aener31ly ox 300 I

comprises for each channel a correlator 302 each cGmDrising a first multiplier 304, a second multiplier 306 and a difference circuit 308. The first multiplier 304 has one input that receives the incoming sequences so and a second input that receives the ; first locally generated reference sequence f hazing a sequence shift that corresponds to eye sequence shift of a redetermined transmitter. The multiplier 306 has one input that receives incoming sequences so and a second input that receives the second reference sequence c having an unassigned equines shift. The outputs of the two multipliers 304 and 306 representing, respectively, the products of the incoming sequences and the to locally generated reference sequences are applied Jo the inputs of difference circuit 308. The difference oddity is a lied by an Lntegra~e and dump type filter 310, watched Jo the period of a bit at the chip rate, Jo develop a signal TV for each channel as follows:
TV 5 strut - e(tN)~dt I

wherein TV and so are analog signals while f and c art binary signals. the ought of the i~teg~te and dump circuit 310 is allied to a sample and hold circuit 312 which monitors and stores the magnitude rid polarity ox the integrator output TV
this value if applied to a conventional microprocessor 314 that in response to outputs prom all N of the detectors 302 extracts tune binary data run the predetermined transmission and develops a timing error signal to retain the receiver locked in synchronism with the predetermined ansmitter, as discussed in more detail below.
The analog multiple channel correlation detector shown in figure 11 requires a substantial number of calibration adjustments associated with the multipliers 304, 306, the difference circuit 308, the integrate and dump circuits 310 and the say to and hold circuits 312. In practice, an 8-channel detector of -is rye requires a~prox~matsly 80 calibration adjustments.

213~3~

If only the polarity ox the reference sequences f and c is used, considerable simDli_ication ox the system results, with only a slight degradation Lo Performance Because the two reference sequences en_ binary (bipolar) signals, multiplication can be achieved in an N channel correlator using I two input analog multiplexes and one invert shown in Figure 12. In this implementation, the binary reference signal determines whether the input Saigon so or an inverted input signal sty ) is selected to be applied to subtraction circuit 308. wearing in mind that the desired output or each of the N difference circuits 308 is so k (~) - c], each channel in the correlation detector 400 shown in Figure 12 comprises a first input multiplexer 402 and a second two-input multiplexer 404 controlled, respectively, by the instantaneous polarities of the first and second Baylor reference sequences I and c. One input of each of the two multiplexes 402, 404 is connected Jo a first line 406 that receives the incoming sequences so and a second input connected to a line 4~8. The line 40~ receives the incoming sequences so inverted Lo polarity by an inverts 410.
The multiplexes 402 and 404 are driven by the reverence sequences f and en) through drivers 412 and 414.
Assuming that the polarities of rut and c ens identical, Roth of the multiplexer 402 and 404 are connected to the line 406~ The input sequence so is thus applied to Roth the positive and negative input terminals of the difference circuit 308 whereby a zero signal is afield to integrate and dump circuit 310 (Fig. 11). If f is positive and eighteen) is negative, multiplexer 402 is connected to line 406 and multiplexer ~04 is connected to line 408. The sequence so is thus plied to the positive input of difference circuit 308 and the inverted swoons so is afield to the negative input terminal of circuit 30Z: the sequence us is 'thus applied to integrate and dump circuit 310.
If, on the other hand, the relative polarities or the Jo reference encase are reversed, tune sequence so is aPDliea to the negative input of difference circuit 308 and the inverted input sequence so is applied to the positive input of difference circuit 308. The signal -sty is thus applied to integrate and dump circuit 310, thereby satisfying the equation S Vet Steinway) k (to) - eighteen) ] -The circuit of Figure 12 is advantageous over the circuit of Figure 11 because analog multiplier calibration adjustments are not required in figure 12, although the inventor 410 requires two (balance and offset calibration adjustments. The number of adjustnen~s required for an eight-channel detector is thus reduced from aFpro~i~ately 90 to 34.
Referring to Figure 13, a further simplification of the circuit shown in figure 11 can be achieved by recognizing what the input Jo each integrate and dump circuit 310 is thy difference between two signals, each Ox which is the input sequence so multiplied by a I or a -1, with the output king zero when the two reference sequences are equal to each other. In accordance with Figure 13, the ON multipliers and the N subtracters are replaced, in circuit 500, by N three-input analog multiplexes 502. One Lout of each of the multiplexes 502 is connected to a line 504 which receives the input sequence so Jo second input of multiplexer 502 is connected to a line 506 thigh receives an inversion so of the input sequence, inverted by 508. The third input of ~wltiplexer 502 is connected to a line lo that in turn US is connected to ground.
The first reference sequence run is connected directly to the control input of multiplexer 502 trough on inverter/driver ;12. Also connected to the control input of multiplexer 502 is 8 exclusive-CR circuit 514 having inputs connected respectively to the two reference sequences run and eighteen).
When the two reference sinuses are equal to each other, the output or the exclusive-OR Circuit 514 drives the multiplexer to line Sly, cays my the output of multiplexer 502 to generate a zero signal Jo ir.t_grate/dumD circuit 310 (Fig. if). If the first ~8~36 reference run equals 1, the output Y of multiplexer 502 else so If rut equals 0, on the other hand, the m~lltipl~xer output Y equals -so. The output of the difference circuit thus generates the signal ~(tn)[r~tn) - eighteen)] and the integrate and dump output for each channel is S Stratton - end as required. (6) o Jo circuits for implementing the three-inFut analog multiplexer 502 ox Figure 13 are Cowan respectively in Figures aye and 14b. on Figure aye, each of the two to input multiplexes 600, 60~ have the following characteristics:
x zoo when A = 0;
x I Al when A = 1.
The first reference sequence f is connected to control terminal A of multiplexer 600 and to one input ox an exclusive-OR circuit 604. The second reference sequence c is connected to a second input of excl~sive-OR circuit 604. The output of the exclusive-OR
604 is connected to the control terminal A ox multiplexer 602.
The I g sequences so are connected to one iota tPrmunal Al of multiplexer 600, and, trough an inventor 606, to the second input Jo of the same multiplexer. The output x of multiplexer 600 is applied to one input Al of multiplexer 602;
the second input I of multiplexer 602 is connected to ground.
The output Y ox the multiplexer shown in Figure aye is defined by the following truth table, which corresponds to the required equation Y Stratton - eighteen)].
TABLE III
.
lo c r 3 e 0 1 1 us 1 0 1 so - 28 ^

In the embcdi¢ent of the three-input multiplexer 606 sown in Figure 14b, the output x is connected selectively to any one of the four inputs Jo, , x2, X3, depending upon the binary values of control inputs A, B. The input sequences so are connected directly to input x2 and through an inventor 608 to input I Inputs Jo and are connected to ground The two reference sequences c and f are connected respectively to control inputs A and B of multiplexer 606.
the operation of multiplexer 606 is described by the truth table set forth above with respect to Figure aye and alto provide s the desired output Y.
m e correlation detector embodiments of Figures 11-14 are based upon the analog technique of integrating a continuous signal. The number of calibration adjustments required can be reduced further by replacing analog integration in the correlation detector by discrete signal summation. Referring to Fissure 15, correlation detector 700, provided in each channel ox the receiver, digitizes the incoming sequences (t) end Algebraically sups the digitized signal in an accumulator over a period of time equal by a bit period. The difference equine the initial and final values in the accumulator represents the value of so integrated over a 'oft period. Accumulation is controller by the values of the reverence sequences f And c. when two two referent ox sequences are equal, the accumulated value it unchanged. When f and c are unequal, the accum~l1ation is incremented or decrement Ed by the value of so depending urn the value of f.
Correlation detector 700 comprises an analog-to-dlgital converter 702 that receives the analog sequence so and in rest nose generates a corres~cndLns digital Saigon it output term m at I. The output of analog-to-digit3l converter 102 is afield to one input A of an ~dder/subtracter circuit 704 having an output a lied to the input of an accumulator c~gister 706.
The output of ye assimilator 706 is acDlied to output register 708 and also to ye second input 3 of adder/subtracter 704.

I

Operation of the units 702-708 as well as of a sequencer 710 ore synchronized to a bit period T. Sequencer 710 in turn controls the conversion times of AND convertor 702 and the accumulation times of accumulator resister 706 at outputs 712 and 714, respectively. The accuowlator register 706 is also controlled by the values of the Jo reference sequences f and c through exclusive OR gate 716 and AND gate aye.
m e ~dder/subtracter 704 develops an output signal which is the sum ox the digitized input sequence so and the Canaanites of accumulator register 706 when reference sequence f is 1 and generates the difference between the accuml1lator register contents and the digitized value ox Lout sicken ox so whey reference sequence f is zero. Selective addition and subtraction ox the TV signals applied at adder/subtracter puts A, B are controlled by the signal applied at input F, developed by reference âequ~nC~
f through an inventor 720.
It f equals c, thy exclusive-OR gate 716 develops a logic 0 signal that i applied to one input of ED gate 718. To the output input of AND gate 718 is a write-accumulation signal de~Jelop~d by sequencer 710. Sequencer 710 alternately deYeloFs a "convert intone signal applied to A/D converter 702 to provide an analog-to-digital conversion of into sequence so and a ''write accumulator" signal which ads or subtract, the instantaneous value of so to the current accumulated value, to ye applied to output register ~08 and when to micro~rccessor 31~ (Figure Lo) which develops binary output and timing error signals.
Thus, the content of the accumulator juster 106 remains unchanged when f equals c under control of an exclusive-OR
gate 716. when f equals a logical, the convent of assimilator register 706 is incarnated by the value of the incoming Seneca so when fit) equals a logic 0, on thy other hand, the convent of the accumulator register is decrement Ed by the value of the input sequence so This has the effect or multiplying so by I 1 or - I and integrating.

I

The correlation detector 700 of Fissure 1; is generalized Unto an N-ch~nnel correlation detector 8Q0 in Figure 16. The reference sequences run and eighteen) are applied to an Input latch 802 having run and eighteen) outputs that are applied respectively to a pair of N to 1 multiplexes 804, 806. The outputs or the two multiplexes 804, 806 in turn are applied Jo the inputs of exclusi~e-CR gate 808 that controls accumulator memory 810 rough AND gate 812.
Accumulator Emory 8L0 in Figure 16 corresponds to accumulator register ~06 in Figure 15~ Memory 810, however, contains a plurality of memory regions corresponding to each channel and addressed by a channel sequencer 814 controlled by the Output of sequencer 816. Similarly, the output of accumulator memory 810 is applied to an output Myra 813 that corresponds to output register ~08 in Figure 15. Emory 818, however, contains a plurality ox me rye regions corresponding to the correlation channels and addressed by the output of sequencer 816.
The incoming sequence sot) is sampled by a sample and hold circuit 820 and aloud to analog-to-digital courter ~22 wherein the incoming analog Seneca so is digitized end æ Dried to aad~r/subtracter 324 in a manner described with respect to Figure 15 .
In operation, sample and hold circuit ~20 samples the incoming analog sequence sot) and converts the samples to corresponding digital values in s~ncnronism with the bit period T
developed by microprocessor 314 (Figure Lo and aloud Jo sequencer 816. The content of the accumulator Emory 810, within each memory region addressed by sequencer 816 is incremented or decrelented by the current value of so depending upon the value of the reruns sequin ox f at the correspondL~g channel. The circuit coo thus successively samples toe input sequence, multiplies the Sweeney ox by I or -1 and integrates or each channel I, ~mder control ox channel sequencer 814 and sequencer 816, us well as of the microprocessor ala. To accumulator memory 810 and output Emory 818 thus monitor N accumSllation channels, with time synchronism of signals during charnel sequencing being preserved by the sample and hold circuit 820 and the input latch 802.
erring row to Figure I another digital implementation of a single channel correlation detector 900 comprises a conventional voltage-to-frequency converter 902 thaw receives the absolute value of input sequence so through an absolute value circuit 904. Absolute value circuit 904 is required because the lug vol~ge-to-~requency converter 902 responds, as is conventional, to a unpiler LnDUt signal. Voltage~to-~req envy converter 902 converts the instantaneous magnitude of the incoming sequence so to a single corresponding frequency signal to key applied to an up/down kismet 906 through one irSput of an AND gate 908.
Ida input sequence so to kiwi applied to an analog comparator 908 which keeps track of the polarity of the iota sequence so In other words, the output ox the analog comparator 908 is representative of the sign of the input sequence so The reference sequences fit) and c are addled to the remanning input of gate go through ex~lusive-C~ gate 910.
The upon counter 9~6 is controlled by a second axclusive-CR gate 912 that receives the output of the analog ccmpara~or 908 end the first no rink sequence f. Thus, the up/down counter is controlled to increment when the signs of the input SeqUeSlCe so and refesenc~ sequence f are the same;
otherwise the counter is caused to decent The output of counter 926 is Allah d to a latch 914 synchronized to bit Period I.
The clock CUR of up/down counter 906 is disabled by exclusive OR gate 910 when the two reference sequences f arid eta are equal to each other. Otherwise, the counter clue k is enabled and the counter 906 tracks the incoming sequence sot). In owner words, when f is l, the counter counts up for a oust polarity sequence bit so and counts down or a nega~i~Je polarity sequence bit so When the reference sequence f is a logic I

32 - , zero, on the other hand, accumulation is subtracted and the count direction is reversed.
The circuit 900 or Figure 17 is generalize to N channels of correlation detection by circuit 1004 in Figure 18. In circuit 1000, voltage-to frequency converter 1002, absolute value circuit 1004 and analog cGm~arator 1006 correspond to corresponding c~m~cnents in Figure 17 and are gammon to all channels. down counter 100~ as well as AND gate 1010 end excl~lsive-OR gates loll and 1014, hcwever7 are duplicated for each channel. The output of each binary up/d~wn counter 1008 is applied to a latch 1016, ccm~only synchronized to a bit period T. the outfought of the N
latches are applied to microprocessor 314 touch as shown in jig.
Lo) which processes the individual channel correlation signals and in response develops binary data recovered from the predetermined l; transmitter and timing signals to shift receiver timing into synchronism with the predetermined transmitter.

dynamic Synchronization As discussed above, static synchroni2aticn involves establishing predetermined delays in the receiver that ~orrespc~d to different proFaga~ion tomes associated with different transmitters. Static delays, preset in the receiver during initial set-up, synchronize the transmitter and receiver to within one cove chip of each other. Perfect correlation it then established by microprocessor 314 in response Jo the Russian signals dev~loF~d by the correlation detectors described above.
~icroerocessor 314 more specifically processes the channel correlation signals to control receives timing to synchronize to the predetermined transmitter in two stages; namely, fine and coarse tuning, followed by synchronization correction, if secessar~, to the proper pulses of the system clock.
Referring gain to Figure 10, it is recalled that code correlation is a function of code chip time delay differences between received code and a reference code and, depending upon 8~3~;

the particular correlation properties of the code employed, has a peak when synahroni~ation is achieved and has an absolute value that dross to zero as the synchronization dif.er~n ox approaches a code chip or greater. Data are recovered from the correlation pattern, buzzed upon the recognition that the sign of the pattern depends upon the data bit used to modulate the transmitter. Thus, when thy receiver and a redetermined transmitter ens properly synchronized to each other, transit data are recovered TV
monitoring the sign or the voltage Al at the prowar correlation channel.

Fine Tuning Referring to Figure 19, a correlation pattern corres~cnding to the correlation pattern shown in Figure 10 is identified my I 1100. This is an "in-phase" correlation pattern, with coarse correction Gunwale Al, V2, V3, V6 and V7 that are used -to determine which of the correlation peaks corresponds to the primary channel, with maximum correlation at synchronization. An additional pair of channels V4, V5 are fine, or vernier, correction channels, which maintain receiver synchronization by maximizing the correlation output of the primary channel Al. In the foregoing discussion, it should be recognized that all reverences to fraction of a code chip are related to the ratio between the carrier 'equines and cede generation frequencies. us one example, the carrier frequency is ;670 I and the cede generation frequency is at 3870 bits/second, 90 that references Jo fractions of a cede chit are related by a ratio of 3/2, allowing three peaks per code clip. The additional correlation curve 1200 in Figure 19 is a quadrature-ehase ccrrela~ion cure that is displaced prom the ions correlation curve by 90 degrees. The significance of the quadrature-phase correlation curve is await the value of the ~uadkature-phase curve is at Roy when the value of thy infuse quadrature curve is at a maxim. AS shall ye discussed below, signal processing, and particularly correlation peak detection, is sL~plified using quadrature-chas2 correlation.

I

ekes then@ are three correlation peaks per code chip, assuming that the primary correlation channel Al it within a code chip ox being properly synchronized, the ~rLmary channel Al is within one-sixth ox a code chip of a "local peak. Fine tuning cause the receiver to adjust its timing, under control ox microprocessor 314, such Nat the correlation channels Al, V2, v3, V6 and V7, spaced apart rum each other by one-third of a code chip, are all located at lock Quick One mucked of adjusting receiver timing to locate the five correlation channels to local peaks is by serial hunt m g shown in the flow ah rut given in Figure aye mix involve use ox a preamble of a length up s), where S it the number of smoothing on each bit and p is equal to one-sixth (in this exile) of a code chip period divided by the receiver correlation resolution, or the number of cocr~lations of minimum resolution requited to adjust the receiver from a synchronization null to a peak.
or each data bit in the preamble, the receiver prLDary correlation channel Al timing it adjusted by a mu us fraction 1/6(p) of a code chip (step 1320) and the magnitude of the correlation voltage Al is stored (1330). This process is repeated until the receiver has clanged its timing owes 3 maximum of a full one-third ox a code chip (1340). m crofter the point at which the magnitude ox the primary correlation Al is at a maximum is selected as being the local weak (1350), and the timing ox the receiver is adjusted to position channel Al at thaw point (1360).
on alternative fine tuning nRthod controlled by microprocessor 314 is the use of fine tuning channels V4 and V5 shown in Figure 19. m e fine tuning channels V4 and Jo provided by an additional pair of correlation detectors (not shown, are offset in time rum the primary correlation channel Al by an equal traction ox a code chip that is less than one-sixth of a code chip. Optionally, a preamble may be included in the method, having a worst case length ox p s with a minimum receiver correction (resolution) icing 1/6(p) of Jo cede kiwi?. Referring to 8~36 Figure 20(b), the correlation voltages V4 and US are applied to MicroPro essayer 314 (step 1950) along with the correlation voltage of tune primary channel AL. By ccm~aring the relative magnitude of TV and V5 (steps 1960, 1970), the microprocessor determines the ; direction toward which receiver timing is to ye shifted (steps 1980, 1990) to position the pry channel Al at the major local correlation peak, system of this type is shown schematically in Figure 21. Prcyr~mming of microprocessor 314 is omitted for brevity, but is considered routine to implement based upon the simplified flow chart of Figure 20(b) and the discussion herein, Another alternative fine tuning method involves the use ox a channel whose timing is generated with a quadrature-phase carrier. Recognizing from Figure 19 that tune nulls of the quadratur~-phase correlation pattern 1200 occur at the peaks of lo in-phase correlation pattern Lowe, an error voltage may ye developed by microprocessor 314 based upon the sign of the product of the ions and quadrat~re-phase patterns. The sign of the error volt go thus indicates a direction to which receiver timing must be shifted to cause the receiver correlation channels to synchronize to local correlation peaks. It is also possible to apply the magnitudes of the in-phase and quadrature-phase correlation voltages isle and 1200 to determine not only the direction of shirt of receiver timing to achieve synchronization but also the amount of shift roared to obtain a local Freak.
Thus, in accordance wit another aspect of toe invention and as summarized in the flow chart of figure aye the in-phase Al and quadrature-phase vlq correlation voltages are measured (stop 2050). The ratio ox the ions Al and ouaarature-phase Vlq correlation voltages is calculated (2060), and if the ratio is poqLtive (2080), the Jo correlations en- resumed to have the same polarity and receiver timing delay is increased (2095);
otherwise, the two correlations are presumed to have opposite polarities and receiver timing delay is decreased (2090). Jo prevent receiver timing from being clanged if the receiver is I .

perfectly synchronized to the predetermined transmitter, and to avoid complications caused by delay in the receiver whereby a correction decision is mode using information that is more than one data bit old, the absolute value of two ratio Vl~Vlq, which is essentially a cotangent junction, is monitored. A table stored in a memory associated with microprocessor 312 relates the ratio Vl~lq to the number of fine tuning corrections, e.g., 1/48~h of a code chip for each correction, to reach optimal synchronization The table is set forth below.
VIE IV
umber of Corrections (dual fractions of a Code Chip) _ _ O Owe 1 5.02 2 2.41 3 1.43 4 1.30 0.66~
6 0.~1~9 7 O.lg99 Thus, the number of corrections applied to receiver timing is US determined directly from Vl~Vlq, and there is a correction dead band when the ratio is greater than 5.02, eliminating receiver huntmg about optimum synchronization. Furthermore, the number ox data bits needed to move the receiver prom a correction null to a correlation peak is reduced from 8 (in this example) to as low as 1, minimizing the length of any required preamble and providing accelerated serial hunting. Finally it it possible to inhibit tracking corrections on consecutive data bit without decreasing the tracking rate of the receiver, thereby eliminating overshoot.

Signal Presence Detection The provision of quadrature-pha~e Vlq as well as Ln-phase Al correlation voltages furthermore makes it possible to determine a signal present within a background ox noise. As summarized in the program wow chart ox Figure 22(b), when only noise is present at the receiver input, both the in-phase V7 and quadrature-~hase Vlq volt will have approximately the same value R, such that the ratio Vl~Vlq will be close to unity. With both signal and noise present, however, fine tuning maximizes Al and minimizes Vlq to obtain a ratio such greater than unity. The cation Vl~Vlq is thus used as an indication of signal present. In practice, the ratio may be monitored over a number of data bits, with smoothing techniques or majority voting being applied to ensure accuracy.
Circuitry for detecting presence of a signal in a background l; of noise is shown in Figure 21~ with microprocessor 314 developing signals Al and V~q Lo response to the outputs of the correlation detector discussed above. The signal Al, Vlq are processed with the microprocessor 314 to develop the ratio VlJVlq and the absolute 0 value VlJVL~ of the resultant is magnitude cared with a predetermined threshold magnitude to determine whether an incoming signal represents a data transn7ission or whether it is merely noise.
Following determination that the receiver is tuned to a local peak U51~9 fine tuning as described above, it becomes necessary to determine through coarse tying, whether the current local peak is the "correct" local perk such that the receiver has jest correlation.

Coarse Tuning o In accordance with one embedment coarse tuning ox the receiver to a predetermined transmitter to ensure that the receiver is tuned to the maximum, and other than a secondary, correlation peak involves serial hunting wherein, having ~ncs idea a point as a local Seiko, the receiver is adjusted in :~Z~3~

multiples of one-third of a code chip to measure the magnitude of the receive signal at each adjacent local peak. Once the magnitudes of the peaks are determined, a decision as to the roper peak is made. because the magnitudes of adjacent peaks near the center of the correlation pattern are difficult to distinguish tram one another due to channel filter distortion, a conventional ~center-of-mass" approach may be used to identity the maximum local peak by basing the decision on the relative values of all channels rather than on only a selection of the channel having the greatest correlation magnitude.
the microprocessor 314 is programmed in a coarse tuning, serial hunt made to cause the receiver, hollowing identlri~ation of a local peak, to shift in timing by multiples of one third of a code chip, measure end store correlation magnitudes and make l; cc~lparisons using the center of maws approach or other-approach to identify the correct correlation peak. Serial hunting requires a transmission preamble of length W s where W is the width of the peak search range yin thirds of a code chip) and s is the number of bits of smoothing in the voltage read mugs.
In Figure 23, a simplified flow chart of programming of microprocessor 314 to provide coarse tuning by serial hunting includes test at step 1200 to determine, using fine tug no as discussed above, whether the receiver is at a local peak. If the receiver is not at a local peak, the receiver is fine tuned until the receiver is determined to be at a local peak. The receiver, once at a local peak, is incriminated (step 1202) until its timing is at N, wherein X is the timing of the local peak obtained during Seine tuning and N it a redetermined number of thirds ox a code chip. The correlation value of R N is measured and stored (step L204), and the receiver timing is ~ecrement~d by one-third of a code chip (stop ~206). The correlation of the receiver and predetermined transmitter is now measured and stored (sued 1208), and receiver timing is tested to determine whether it is (R - I), that is, at the opposite wide o. the initially detected I

local peak step 1210). If not, the receiver timing is again decrenented and the correlation is measured and store.
Otherwise, all the stored correlations are tested (step 1212) to identify a weak correlation.
In accordance with another embodiment, to reduce the Preamble Length, multiple secondary receiver channels, offset from each other by multiples of one-third of a code chit on Roth sides of the primary channel Al Delco primary and secondary correlation signals to lo applied to microprocessor 314. The microprocessor 314 Lo programmed, using center of mass analysis or other analysis, to identify the primary charnel Al which has the greatest maximum correlation and the skinnier channels. By using a multiple number of receiver channels or correlation detectors, rather than serial hunting circuitry or programming the length of l; the ruble required for coarse corrections may ye reduced to the number of bits of smoothing, s. this assumes ox course that or the desired width of starch, a channel exists with common offsets of multiples of one-third of a code clip on Roth sides of the primary correlation channel Al.
With multiple receivers it is not necessary to program the microprocessor to serially hunt. m e microprocessor 314 it instead programmed to simply compare the outputs of the correlation detectors, all tuned to a local peak, to identify the peak having the greatest magnitude.
2;
Timing Signal Correction If the data bit rate of he transmission is less than one-half the pulse repetition rate of the timing source, the transmitter and receiver may become locked to different timing pulses even though they appear to be perfectly synchronized to each other. or example, for a data bit rate of 30 bits per second, a liming pulse source of 60 I and a carrier frequency located between 60 I harmonics, the transmitter Jay become locked to a lust 60 I timing pulse with the receiver locked to toe next 8~36 successive 60 I timing pulse. An alternating data transmission will not be detected due to improper receiver data timing recovery with otherwise perfect synchronization between the receiver and transmitter.
To illustrate this condition more clearly, Figure aye) is a diagram representing the timing pulses to which the receiver and a predetermmed transmitter are synchronized. The transmitter carrier tat shown in Figure 24~b) and transmitted data re~resentmg alternate ones and Zeros are shown in Figure 24~b). assuming that the receiver and transmitter are synchronized to the same tuning pt~l5e5, the integrate and dump circuits 310 of the receiver will be synchronized to the transmitted data inversions so a to dump at the trailing edge of each datum, as shown in Figure 24~d), where "dot" designate integration duo points. The sampled l; integrator output it thus a replica of the data embedded within the ~ransnission~
If the transmitter and receiver en_ not synchronized to the same timing pulse, however, the integrate and dump circuits 310 will not be properly synchronized to the data being transmitted.
This condition is shown is Figure 24~c~, where the integration dump joints occur between transmission data inversions, and the sampled output of the integrator 310 is at zero.
In other words, with the recover and transmitter respectively synchronized to successive, rather than the same, timing Pulses, it is impossible to recover any of the transmission date. It is therefore necessary to test the receiver and transmitter to ensure that the two unit are synchronized to the same, rather than successive, timing pulses.
In accordance with one aspect of toe invention, associated with the primary receiver channel Al is a secondary receiver channel Al' having a ouilt-in additional delay of one-half a data bit. One of the two channels Al and Al' will always therefore detect the transmitted signal. detsrnination is made by applying an alternating date preamble associated with the ;~2Z&1~

transmission to the prowar and secondary receiver channels. my camper my the magnitudes of the correlation outputs of the two receiver channels, the correct channel (having the larger correlation magnitude) is thy one synchronized to the same timing pulse as the transmitter is. Data are monitored at the "correct"
channel only.
A simplified circuit or synchronizing receiver timing to cause the receiver and transmitter to ye locked to the same timing pulses as shown in Figure 25. microprocessor 314 develops a I secondary channel ~71~ offset from channel V1 by one-half ox a data bit. In resFcnse to an incoming sequence having an alternating preamble, the microprocessor compares the magnitudes or the data output iron the channel Al and its half bit delayed channel I
and idiots the one channel having the larger magnitude This Lo channel is thus presumed to be the one which is locked to the same timing pulses as the transmitter is, and is reapplied to the microprocessor for data recovery.
In an alternative elbodi~ent of ye invention, the nest for the secondary receiver channel Al' may key eliminated. The transmitter and receiver can ye synchronized when the Tony reference frequency is less than or equal to the data sampling rate and the rails of the data sample my rate to two timing reference frequency is an integer by combining more than one ox the consecutive data sales together Jo yield one data point or bit. my combining these data samples, an optimum data sample point may by determined while receiving an alternating sign preamble by comparing the magnitudes of all possible summations and selecting the sample which give a maximum output. If each sample is assigned to its own synchronization point, then synchronization may be accomplished Joy locking to the tire that juicy the maximum output.
For example, if the timing signal has a frank of 60 Hertz and a data sampling rate of 30 samples per second, or a dot rate of 30 bits per second each data sample is used to yield one data I 51~3~;

- I -point or bit. For data rats of 15, 7.; or 3.75 bits per second two, four and eight consecutive data aimless are used to yield one data bit. In addition to eliminating the need for a redundant data channel, the above technique eliminates the need for tune data sampling rate to be the same as the data rate. In fact, sampling may occur at a rate higher than the data fate. This allows the data samples to ox combined digitally, for example in a mlcropcoce~sor, and allows the data rate to be independent of the actual hardware timing.
Oat Recovery Data recovery in spread spectrum cyst my is well 'Nina us background, referee is made to section 5.3 of the Nixon text mentioned earlier, and particularly to the discussion of Kowtows loo demodulators beginning on page 15;.
Because the spread spectrum system as provided here m includes multiple correlation channels, data recovery it improved in accordance wit h one aspect of the invention by extracting data at each channel rather than at only a single correlation channel.
It is thereby possible to lower system message error rate and possibly to also reduce the length of or eliminate any required preambles or receiver synchronization.
With reference again to Fig~rQ 19, it it noted that the correlation pattern 1000 is centered about the primary correlation channel Al. The sign ox the primary correlation channel Al is dependent upon the sign of tune data being transmitted. A positive value of Al thus corresponds to a logic 1 being transmitted Norris a negative value of tune correlation Al corresponds to a logic 0 being transmitted.
The correlations at V2, V3, V6 and V7 also have values that correspond to the sign of the data being transited.
Specifically, the relationship of tune voltage outs at channels Al, V2, V3, V6 and V7, in the absence of noise and distortion, are descried as follows:

AYE

V2 - V3 - Al- Al V6 V7 - R2~ Al where Pi -2/3 In accordance with the Invention, the data sign at the output ox each correlation detector, following proper receiver synchronization, is monitored. Depending upon the characteristics of noise and distortion, data may ye extracted using only the outputs at channels Al, TV and V3, with an effective signal-to-noise ratio gain ox lull us Rj)2 8 j-L ( ) lo (l+ V L) us 2 + 2 (~1+1/L) us (u2+u3) 2 (R2+1jL) us us i-l where Rj 2 relative noise free amplitude of Vj with respect to "1' i = 2, 3 (Kj Y Al in the distortion free case), L = the length of the ps~udo-candom code and Us weighting factor or Vj, j = 1, 2, 3. The weighting factors are selected according Jo the particular distortion present.
Figure 26 it a simplified circuit diagram showing microprocessor 314 responsive to channels Al, V2 and V3 and programmed to combine all three correlation channel outputs to extract transmission data, with weighting factors selected according to particular distortion known to be present on the transmission medium. table V illustrates the signal-to-noise enhancements under a few possible distortion and weighting factor scenarios.

~.~Z~36 my ON
r~EIG8TING FUSSERS FOR Vj DISTRAIN F~CIOFS FOR Vj CEMENT
ulU2 us Al I k3 F~CIOR
. _ _ ,_ _ _ 1 -1 -1 1 -1 -1 1.
_ _ , _ , __ 1 -1 -1 1 -0.9 -0.9 1.25 _ . _ . _ 1 1 -1 1 -0.8 -ova 1.082 _. _ _._ _ 1 -1 -1 1 -0.7 -0.7 O.g22 , _ Jo _ _ 1 -0.34 _0.34 1 -0.67 -0.67 ~.971 . _ _ . _ . . I
1 -0.67 -0.67 1 -0.67 -0.67 00918 ._~ _ . . Jo _ .
1 -0.9 -0.6 -0.9 I 1.055 _ __ . . _ 1 -0.8 -0.8 1 -0.8 -0.8 1.09 . ,., _ . . _ _ 1 -0.9 -0.9 1 -0.9 -0.9 1~252 ___ _ _ _ .
An additional advantage of providing a recovery on all channels of the receiver is that random and burst errors Rich tend to act all channels, can ye identified and ignored. This is similar to signal presence detection us my in-phase and q~adrature-~hase correlation outputs, as discussed above, but employs all channels rather than orthogonal outputs associated with a single channel.
Furthermore, as an additional advantage of obta m in data recovery at alp correlation channels or at lookout several correlation channels, it is w Sybil to monitor synchronization during message reception. Although synchronization adjustments are not feasible during message reception, the message content may ye recovered, without rspsats, using the additional receiver channels.
In this disclosure, there is shown and described only toe referred embodiments of eye invention: however, it is to be ~2~3~L36 understood that the invention is capable of use in various other combinations and environments and is capable of changes or modifications within the scope of the inventive concept as expressed herein.
s

Claims

WHAT IS CLAIMED IS:

1. A direct sequence spread spectrum code division multiplex system, comprising:
a timing signal source;
a plurality of transmitters synchronized to the timing signal source and each transmitting data spread by a bipolar pseudo-random code which is a different assigned shift of a common bipolar code sequence; and characterized by:
a receiver synchronized to said timing signal source for receiving said data signals and discriminating the signal transmitted by a predetermined transmitter spread by a bipolar pseudo-random code having a predetermined assigned code sequence shift from signals transmitted by the other transmitter, the receiver including means for generating a first bipolar pseudo-random code that is a replica of the transmitted common bipolar pseudo-random code and has the predetermined assigned code sequence shift, means for generating a second bipolar pseudo-random code that is a replica of the transmitted common bipolar pseudo-random code and has an unassigned code sequence shift, means for processing said first and second bipolar pseudo-random codes to obtain a trinary code sequence and means for cross-correlating said transmitted signals and said trinary sequence.

2. The system of claim 1, wherein said bipolar code sequence is a maximal length (ML) sequence.

3. The system of claim 1 or 2, wherein said bipolar code sequence is a phase shift keyed signal.

4. In a direct sequence spread spectrum code division multiplex system including a plurality of transmitters synchronized to a common timing signal and each transmitting a data signal s(t) spread by a bipolar pseudo-random code which is a different assigned shift of a common bipolar code sequence:

a receiver synchronized to said timing signal for receiving said transmitted signal spread by a bipolar pseudo-random code having a predetermined assigned code sequence shift, and characterized by:
means for generating a first bipolar pseudo-random code that is a replica of the transmitted common bipolar pseudo-random code and has the predetermined assigned code sequence shift, means for generating a second bipolar pseudo-random code that is a replica of the transmitted common bipolar pseudo-random code and has an unassigned code sequence shift, means for processing said first and second bipolar pseudo-random codes to obtain a trinary sequence and means for cross-correlating said transmitted signal and said trinary sequence.

5. In a direct sequence spread spectrum code division multiplex system comprising:
a timing signal source;
a plurality of transmitters synchronized to the timing signal source and each transmitting a data signal spread by common bipolar pseudo-random code which is a different assigned shift of a common bipolar pseudo-random code sequence: and a receiver synchronized to the timing signal source for receiving said transmitted bipolar pseudo random code having a predetermined assigned code sequence shift:
a method of synchronizing the receiver to a predetermined one of said transmitters transmitting a data signal spread by the bipolar pseudo-random code having said predetermined assigned code sequence, characterized by the steps of:
generating a first bipolar pseudo-random code that is a replica of the transmitted bipolar pseudo-random code having said predetermined assigned code sequence shift; generating a second bipolar pseudo-random code that is a replica of the transmitted bipolar pseudo-random code having an unassigned code sequence shift; combining said first and second bipolar pseudo-random codes to obtain a trinary sequence; cross-correlating said transmitted binary pseudo-random codes and said trinary code sequence; and in response, generating a receiver timing signal.

5. The method of claim 5, wherein said processing step includes a subtraction.

7. The method of claim 5, wherein said binary code sequence is a maximal length (ML) sequence.

8. The method of claim 5, 6, or 7 wherein said binary code sequence is a phase shift keyed (PSR) signal.