CA2044166C - Digital receiver for spread-spectrum signals - Google Patents
Digital receiver for spread-spectrum signalsInfo
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- CA2044166C CA2044166C CA002044166A CA2044166A CA2044166C CA 2044166 C CA2044166 C CA 2044166C CA 002044166 A CA002044166 A CA 002044166A CA 2044166 A CA2044166 A CA 2044166A CA 2044166 C CA2044166 C CA 2044166C
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- correlator
- receiver according
- digital
- correlation
- stages
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
- H04B1/709—Correlator structure
Abstract
The receiver contains a correlator (TIC) for the correlation of the received signal with a reference code (R(k)) and a digital signal processor for evaluating the results of the correlation. The correlator is constructed as digital, time-integrating, multistage correlator (TIC), the individual stages (K1 to KN) of which are connected on the one hand to a digital delay line (17) fed with the reference code (R(k)) and to which on the other hand a digital signal (S(k)) recovered from the received signal is applied, the length (D) of the delay line (17) being adapted to the maximum expected impulse response of the transmission channel.
Description
2~4~
Ascom Zelcom AG, CH-8634 Hom~rachtikon PA-10/175 Diqital receiver ~or spread-spectrum siqnals The present invention relates to a digital re-ceiver ~or spread-system signals generated by multiplica-tion of an information-carrying signal by an auxiliary function, having a correlator for the correlation of the received signal with a reference code, and having a digital signal processor for evaluating the results of the correlation.
European Patent Application No. 89,117,388.2 describes a digital radio transmission system which uses the spread-spectrum ~echnique in which digital signal proces-sing is used but the correlators of the recei~ers are constructed discretely with analog components and are thus relati~ely costly in terms of space and power re-quirements. In addition, the analog components have the known drifting and aging problems and they also restrict the applica~ion possibilities of the digital signal processing.
By means of the invention, the implementation of digital correlators with sufficiently high time-bandwidth products will now be made possible, so that receivers can be built which are compact and economical in terms of power consumption and which do not have any drifting and aging problems.
This object is achieved according to the inven-tion by means of a digital, tLme-integrating, multistage correlator, the individual stages of which are connected on the one hand to a digital delay line fed with the reference code and to which on the other hand a digital signal recovered from the received signal is applied, the length of the delay line being adapted to the ~imllm expected length of the impulse response of the transmisson channel.
The correlator according to the invention can be integrated monolithically and is therefore economical in terms of space and power and is easy to install. The receivers equipped with correlators of this kind do not have the drifting and aging problems of the kno~n analog 20~
solutions and are extremely flexible. The receivers are particularly suitable for radio tr~n~mi~sion systems with multipath propagation, but they also permit multiple use of the transmission channel by means of multiple access (code division multiple access = CDMA). Such receivers are used advantageously for radio systems with microcell arrangement in locally ].imited areas such as factory premises, houses or multistorey buildings.
The invention is explained in greater detail below with reference to an exemplary embodiment and the drawings, in which:
Fig. 1 shows a block circuit diagram of a digital multipath receiver for spread-spectrum signals, Fig. 2 shows a block circuit diagram of the correlator of the receiver of Fig. l; and Figs. 3,4 are diagrams for the purpose of func~ional explanation.
The multipath receiver according to the invention is a digital receiverfor spread-spectrum signals according 2C to the so-called direct sequence meth~d and it is par-ticularly suitable for radio tr~n~mi~sion systems on channels with multipath propagation. A system of this kind is described in European Patent Application No.
89,117,3R8.2 by the Ascom Zelcom AG, reference being hereby e~plicitly made to the disclosure of said patent application. The digital receiver described below con-stitutes an impLove.~ent of the correlation receiver described in this patent application, the Lmprovement concerning primarily the employed correlators of the receiver, in that now digital correlators are proposed.
Fig. 1 shows a block diagram of a digital multi-path receiver which consists according to the illustra-tion of three blocks, a converter 1, a so-called I/Q processor 2 and of a digital stage 3. In the converter 1, the received broad-band signal which arrives at the aerial of the receiver via paths Pl to Px is filtered in a band-pass filter 4 in order to suppress signals outside the employed frequency band. The output signal of the band-pass filter 4 is amplified in an amplifier 5 and mixed in a mixer 6, at whose other input a local oscillator 7 is connected, to an inter-mediate frequency fIF This intermediate frequency signal now passes into the I/Q processor 2 where it is filtered in a further band-pass filter 8 and then mixed into the base band by multiplication (mixer 9, 9') by a cosine or sine signal of the frequency fo from a local oscillator 10. As a result, two resulting signals I(t) and Qtt) arise which are freed from high-frequency signal portions by low-pass filters 11, 11' and amplified in amplifiers 12, 12'. Subsequen~ly, the two signals are quantized in analog-to~digital converters 13, 13' with a sampling rate of fs = c/Tc and represented as digital values I(k) and Q(k) by b bits, b being for example equal to 8.
In the formula for the sampling rate fs, c desig-nates the number of sampling values per code element (=chip) and Tc the duration of a chip. This duration corresponds to the smallest square-wave pulse length of the auxiliary function used for band spreading, the characteristic of ~0 which of course is known to the receiver in a spread-spec-trum system and which is formed in the receiver by the reference code.
The third block of the receiver, the digital stage 3 which processes exclusively digital signals, consists ; essentially of correlators TIC, of a reference code gener-ator 14 and of a digital signal processor 15. The sequen-ces I(k) and Q(k) are each fed into a correlator TIC and correlated ~here with the reference code R(k).
ence code R(k). The correlation results CI~m) and CQ(m) are read by the digital signal processor 15 and processed further. The latter derives control signals PD from them for controlling the reference code generator 14 and the correlators TIC, calculates the deviation between the carrier frequency fIFand the local oscillator frequency fo and carries out the coherent demodulation of the transferred information bits. The reference code genera-tor 14 supplies the correlators TIC with the reference code R(k), the sampling rate fs and a synchronization signal SY, and supplies the analog-to-digital converters '' ' ' ' '' ' '., ' . .
_ 4 _ 2~
13, 13' with the sampling rake fs.
The requir~ments on digital correlators are det~rmined by the parameters of the in-house radio transmission system. A clock rate (chip rate) for the pseudo-random auxiliary function in the region of 10 to 30 ~z can ba derived from the coherence bandwidth of the propagation channel. In order to permit a sufficiently large number of users in a cellular system with multiple access (CDMA)spreading factors of at least 255 to approxi-mately 4000 are required. Since the synchronization of the local reference code to the received code should also occur digitally in a digital correlation receiver and in this procass each chip must be samplsd at least twice, sampling and processing rates for the correlator of 20 to 60 MHz are obtained.
Since a plurality of signals are superimposed on one another at the reception site due to multipath propa-gation and multiple occupation of the channel by means of CDMA, at the receiver input the signal no longer has a constant signal envelope and must therefore be amplitude-quantized in front of the correlator with resolution of sQ~7eral bits. If this does not occur, intolerable losses ~hen arise. The reference code can, however, be present as a binary signal.
A digital correlator which fulfils these require-ments can be realized as programmable transverse filter or as time-integrating correlator. ~n implementation as programmable transverse filter is known from the publica-tion ~Digital SOS-MOS Correlator: Basic System Component in Experimental Army Spread Spectrum Radio" by N.A. Saethe -cn, B. Skeie and S. Prytz in 2nd Int. Conf.
on the Impact of High Speed and VLSI Technology on Comm.
Systems, London 1983. However! his implementation is extremely costly since with two sampling values per code chip per signal bit at least 2.L memory cells for refer-ence code and signal and 2.L multipliers are xequired (L=code length). The dat~ rate of the correla-tion result is the same as the sampling rate of the input signal and therefore very high. This permits a rapid ~ 5 ~ 2~ 6 synchronization but is unfavourable for the further digital signal processing.
In contrast to the programmable transversal filter, the output data rate in a tlme-integrating correlator is reduced in relation to the sampling rate ~t the input by the number of sl ?A correlation products. With integration over a complete code period, the correlation results only occur - with the information bit rate, which signifies a reduction in the sampling rate of the input signal by the product c.L. (c=number of sampling values per code chip). This reduced data rate can now be easily processed further in a digital signal processor.
From the literature, implementations of time-integrating correlators in CCD technology (CCD: charge-coupled device) are known (B.E. Bur~e, D . L . Smythe: "ACCD Time-Integrating Correla~or", IEEE J. of Solid State Circuits, SC-18, Dec. 1983) and as acousto-optical com-ponents (F.B. Rotz: ~TLme-Integrating Optical Correla-tor~, Proc SPIE, Vol. 202, 1979). Since, on the one hand, the CCD solution is limited to clock rates of a -x;
of 20 MHz and has a dynamic range restricted by offset voltages and clock crosstalk, and, on the other hand, the acousto-optical solution cannot be built up monolithi-cally and is moreover complicated and expensive, these known implementations are not suitable for the digital multipath receiver of Fig. 1.
Although the digi~al correlator TIC is a time-integrating correlator, it is neither a charge-coupled nor an acousto-optical component, but rather its archi-tecture is especially adapted to the requirements of amultipath receiver according to the scanner principle.
The solution found for this which is described below is monolithically integrateable and therefore sLmple to use and it is space-saving and economical in terms of power. The programmable functions of the tLme-integrating correlator TIC and the evaluation of the correlation results in the digital signal processor 15 permit the construction of a very flexible spread~spectrum receiver.
Fig. 2 shows a block diagram of the time-. . .
. , ': .
.
~ .
_ 6 - ~ 6 integrating correlators TIC employed in the rec~iver o~
Fig. 1. The number sequences, designated in Fig. 1 by I'k) and Q(k), of the I/Q processor 2 are designated here in general as representative of both number sequences with S(k). According to the illustration, the correlator consists of N correlator stages Kn, of a control logic 16 and of a digital delay line 17, to which the correlator stages are connected in parallel.
The architectur~ of the time-integrating cor-relator is adapted to the expected impulse response of the propagation channel. This ir~ulse response is, as measure-ments have shown, significantly shorter than the data bit length, so that significant correlation values only arise during a short part of the code length. In the synchroni-~ed state, it is therefore sufficient to correlate only a small section of the complete code with the received signal. For this purpose, the binary reference code R(k) supplied by the reference code generator 14 (Fig. 1) is fed into the digital delay line 17, which has a certain length D, in the present case D=32, and which is operated with the clock fs, for example fs=2/Tc. The reference code is delayed hy the time Td between each of the t~rmin~ls of two successive correlator stages Kn, in which case preferably Td=Tc/2. The n-th correlator stage therefore receives the reference code delayed by n.Td as a re~erence signal.The di~itized output signal S(k) of the I/Q processor 2 (Fig. 1) having, for example, 8-bit resolution is connected in parallel to all corre-lator stages Kn.
Each correlator sta~e ~n contains, as illustra-ted, an accumul~tor 18 and a result memory 19 which are both controlled by the control logic 16, and calcu-lates the product of S(k) and R(k-n). These products are sl ~~ and stored in the respective accumulutor 18 35 over a complete code length k (k=lc times L). At the end of the summation, the correlation value Cn(m) is transmitted into the result memory 19 by a control signal RS' of the control logic 16, and then the accurnula~or 18 is set to zero again with RS. Each result memory 19 _ 7 _ 2~
supplies its correlation value Cn(m) after a correspond-ing control signal REn of the control logic 16 to the digital signal processor 15 (Fig. 1~ which further pro-cesses the results of the preceding correlation during the next summa~ian period. On the basis of the signal REn (Read Enable) and under the control of the digital signal processor in each case a single correlation is read out via a bus.
The design of the time-integrating correlator TIC
can be further simplified by means of a selection circuit 20 which connects a delay~d reference sequence for each correlation stage Kn with a delay which can be programmed by the digital signal processor lS. This simplification is based on the following consideration: the length D of the delay line 17 must be adapted to the ~ximll~ expected length of the channel impulse response which is approxim-ately 1 microsecond in the present case. However, in practice this impulse re~onse always consists of a plural-ity of discrete signal portions with specific delays tp, so that only these discrete signal proportions, there-fore, need to be correlated. For this purpose, correlator stages are placed only at those points tp where signal portions are effectively present. As a result, the number N of correlator stages can be kept substantially smaller than the length D of the delay line 17. In the present exemplary embodiment where D=32, a number of N=8 cor-relator stages is still sufficient. A precondition for this mode of operation is a free progralmmability of the delay of each correlator stage. The programming occurs through the digltal signal processor 15 which drives a line DL via control logic 16, -that programs the ~lection circuit 20 which, itself, conn~cts the reference sequence with the programmed delay to the resp~ctive correlator stage.
In addition to the already mentioned ad~antages of the digital multipath receiver, a further essential advantage consists in the f~ct that the essential func tions of a spread-spectrum receiver can be completely carried out by corresponding operations in the digital - 8 - ~0~ 6 signal processor. The receiver can be characterized by the following states:
- coarse synchronization (acquisition) - operation - re~synchronization on loss of the code synchronism.
As already mentioned, the duration of the irnpulse response the transmission channel is only a frhction of the data bit length and thus of the code length L. For the acquisition, the N correlator stages of the tLme-integrating correlator are now programmed in such a way that the delav positions be-tween succeeding stages differ by a constant delay e.Tc, e = 1 being selec-ted for example. By shifting the receive-side reference code by N.e.Tc after every accumulation ~eriod, t'.1~ complete code length is correlated sequentially with the received signal. The acquisition time Tacq is then Tacq=L.Tc.(L/N.e) in com-parison ~o L.Tc L for a correlator of the type of a pro-grammable transverse filterO The respective reception energy E~m) for each delay can be calculated from the correlation values CI(m), CQ(m) obtained in this way:
E(m)=CItm)2+CQ(m) 2 After searching the complete code length, the position is determined where the m~i reception energy has occurred. In order to confirm whether the energy -~i has really been found, it is e~ ;ned whether the energy around this -x; is grea~er by a specific amount than the noise energy averaged over the complete code length. If this is the case, the reference code generator 14 (Fis. 1) is programmed in such a way that the energy -Yi lies in the centre of the reception window covered by the correlator. Thus, the acquisition is ter~in~ted and the receiver goes into the normal operating state; otherwise the acquisition is repeated.
The normal operating state comprises the follow-ing functions: monitoring the channel ~ulse response (scanning); tracking the phase of the local reference code generator; estimating the carrier phase and coherent or differential demodulation of the signals of the individual reception paths; deriving the weighting _ 9 _ Z0~ i6 functions for the individual reception paths from the channel lmpulse response combining the individual paths and detecting the transferred data bit. If an error protection coding has occurred in the transmitter, the receiver can additionally supply a quality criterion for the following error decoder for ~he so-called soft decision.
The advantages of a multipath receiver then have the m~i effect if a good estimation of the channel impulse response can be carried out. This occurs in the present digital receiver by means of the scann; ng algo-rithm, for which S of ~he total N correla~or stages of the time-integrating corrPlator TIC can be used. The other correlator stages (number = N-S) are required in parallel to the latter for the data demodulation. With the scanner-correlation stages mutually ~nifted by Td, L.Tc correlation values are calculated during a code period. Then, the correlator stages are shifted by S.Td, and ~he calculation of the correlation values is repeated for this position. After ts=D/S code periods, the entire window is searched and the scanning process begins again. This scanning function is illustra-ted in Fig. 3a.
As can be seen in Fig. 3b, the correlator stages required for the demodulation are programmed at the points where the greatest reception power is to be axpected. These correlation values CI(m) and CQ(m) can be interpreted ~ccording to Fig.4a as coordinates of a data vector d(m) = (CI(m), CQ(m)) in the complex plane. Fig.
4a therefore constitutes the absolute value of the channel impulse response measured during the scanning ~unction.
That the measurements of the channel impulse response are significant when scAnning, the scan time ts must be smaller than the i n i change time of the transmission channel. ~he more correlator stages are used for the scAnning, the quicker the changing channels which can be monitored. Hence, correspondingly fewer correlator - ' . , .
stages are then available for the data demodulation. For typical in-house channels in the described system for example S=4 scanner channels are required. Since the time-integrating correlator TIC can be programmed by the digital signal processor 15 (Fig. 1), the partition into scanner and demodulator channels can also occur adaptively.
With the aid o~ the correlation values of the scanner channels, it is monitored whether the correlation 10 window of the time-integrating correlator TIC is cor-rectly positioned in relation to the received signal.
Because the channelimpulse response is usually shorter than the window width, a nois~ power can be detç ;ned from the values outside the impulse response. If the ratio 15 of the power of all the reception paths to this noise power falls below a predete ;ned threshold, a re-synchronization is initiated.
The fine synchronization (tracking~ of the reference code can occur in such a way that the weighted 20 average of the D correlation values, that is to say their median point, is positioned in the centre of the window of the time-integrating correlator. A different method consists in positioning the strongest reception path in each case at a specific point, for example at D.Td/3.
As has already been mentioned, the correlation values CI(m) and CQ(m) of the scanner channels can be interpreted coordinates of a data vector d(m) = (CI~m), CQ(m)) in the complex plane (Fig. 4a). The angle Phi between the data vector and the real axis then corres-30 ponds to the phase shift between the carrier of the received signal and the local oscillator. If the frequencies of the received signal and of the local oscillator coincide (fIF=fo), the angle Phi will remain constant on average and as~ume an average value and only 35 fluctuate around this average value due to the noise in the received signal.
For an optimum detection, by averaging over a plurality of reception vectors, the unit vector e=(xl, yl) is calculated for which the following applies:
.
.~
:
yl/xl=tan(phi) Z~ 6G
xl2+ylz=l Each received data vec~or d(m) is multiplied by the complex conjuga~e vector e:
S B(m)-Re(e .d(m)) From the real part of the product, the transmit-ted data bit is deterrin~d from the sign of B(m); the magnitude of the product is a measure for the reliability of the decision and can be supplied as quality informa-tion to a following error decoder.
If the frequencies of received signal and localoscillator are different, successive vectors d(m) and d(m+l) are rotated towards one another (Fig. 4b)o This rotation Psi is proportional to the frequency difference df and to the bit length Tb:
Psi=2.Pi.df.Tb In this case, in addition to the initial phase Phi, the frequency offset must also be estimated by means of the phase rot~tion Psi. From this, a re~erence vector r can be calculated which now no longer possesses a con-stant phase Phi, but rather rotates with the angular speed 2.Pi.df:
r(m)=(x2, y2) y2/x2=tan(Phi~2.PiOdf.m.Tb)~5 The demodulation occurs in the same way as before:
B(m)-Re(r(m) .d(m)) The estimation of the frequency difference df can occur, for example, by means of a Fast Fourier transform (FFT) by means of 2M successive data vectors d(m).
(m=1... 2M, to example m=1... 32).
In a multipath receiver, each reception path n usually has a different phase shift Phi in relation to the local oscilla~or because of the different lengths of the propagation paths. The frequenc~ difference df is, however~ almost identical in in-house channels fvr all paths, since Doppler effect can be neglected. Therefore, the estimation of the frequency difference for all paths can occur jointly, whilst the phase for each path must be calculated individually. A simple solution is - 12 - 20~
offered here by the differential demodulation of two successive data vec~ors d(m)-1) and d(m). Here, the first vector d(m-1) is rotated by the amount Psi=2.Pi.df.Tb and then used as reference for ~he demodula~ion of the next data vector d(m):
r(m)=d(m-l).e The demodulated bit value is:
B(m)=Re(r(m) .d(m)) The demodulated bit values B(m) of n channels are present in the multipath receiver. Said values have to be combined in an appropriate manner in order to enable a decision relating to the received bit to be made. For this purpose, for each reception channel the expected value of the signal amplitude Gn=[(dn(m)-dn(m)*)~] is calculated as weighting. The bit values B(m) are multi~
plied by the square of Gn and the produc~s are summed over all values of n to form S(m). ~he sign of S(m) yi~lds the value of the detected bit and the magnitude of S(m) is a measure of the strength of the received signal and therefore also of the reliability of this decision.
If the code synchronism is lost in operation, a resynchronization is performed. In contrast to the acguisition, however, the complete code length is not searched, but rather one of the search algorithms known from the literature is used, for example that according to the publication ~Performance Analysis for the Expand-ing Search PN Acquisition Algorithm" by W.R. Braun, IEEE
Trans. Comm., COM-30, 1982.
Thus, all the essential functions of the mul~i-path receiver, in particular the evaluation of theresults of the correlation have been carried out in the digital signal processor, which, in con~unction with the programmability of the functions of the time-integrating correlator, gives the described multipath receiver a considerable flexibility. In addition, the use o~ the time-integrating correlator with the described architec-ture permits a monolithically integrateable solution which is easy to use, space-saving and economical in terms of power.
Ascom Zelcom AG, CH-8634 Hom~rachtikon PA-10/175 Diqital receiver ~or spread-spectrum siqnals The present invention relates to a digital re-ceiver ~or spread-system signals generated by multiplica-tion of an information-carrying signal by an auxiliary function, having a correlator for the correlation of the received signal with a reference code, and having a digital signal processor for evaluating the results of the correlation.
European Patent Application No. 89,117,388.2 describes a digital radio transmission system which uses the spread-spectrum ~echnique in which digital signal proces-sing is used but the correlators of the recei~ers are constructed discretely with analog components and are thus relati~ely costly in terms of space and power re-quirements. In addition, the analog components have the known drifting and aging problems and they also restrict the applica~ion possibilities of the digital signal processing.
By means of the invention, the implementation of digital correlators with sufficiently high time-bandwidth products will now be made possible, so that receivers can be built which are compact and economical in terms of power consumption and which do not have any drifting and aging problems.
This object is achieved according to the inven-tion by means of a digital, tLme-integrating, multistage correlator, the individual stages of which are connected on the one hand to a digital delay line fed with the reference code and to which on the other hand a digital signal recovered from the received signal is applied, the length of the delay line being adapted to the ~imllm expected length of the impulse response of the transmisson channel.
The correlator according to the invention can be integrated monolithically and is therefore economical in terms of space and power and is easy to install. The receivers equipped with correlators of this kind do not have the drifting and aging problems of the kno~n analog 20~
solutions and are extremely flexible. The receivers are particularly suitable for radio tr~n~mi~sion systems with multipath propagation, but they also permit multiple use of the transmission channel by means of multiple access (code division multiple access = CDMA). Such receivers are used advantageously for radio systems with microcell arrangement in locally ].imited areas such as factory premises, houses or multistorey buildings.
The invention is explained in greater detail below with reference to an exemplary embodiment and the drawings, in which:
Fig. 1 shows a block circuit diagram of a digital multipath receiver for spread-spectrum signals, Fig. 2 shows a block circuit diagram of the correlator of the receiver of Fig. l; and Figs. 3,4 are diagrams for the purpose of func~ional explanation.
The multipath receiver according to the invention is a digital receiverfor spread-spectrum signals according 2C to the so-called direct sequence meth~d and it is par-ticularly suitable for radio tr~n~mi~sion systems on channels with multipath propagation. A system of this kind is described in European Patent Application No.
89,117,3R8.2 by the Ascom Zelcom AG, reference being hereby e~plicitly made to the disclosure of said patent application. The digital receiver described below con-stitutes an impLove.~ent of the correlation receiver described in this patent application, the Lmprovement concerning primarily the employed correlators of the receiver, in that now digital correlators are proposed.
Fig. 1 shows a block diagram of a digital multi-path receiver which consists according to the illustra-tion of three blocks, a converter 1, a so-called I/Q processor 2 and of a digital stage 3. In the converter 1, the received broad-band signal which arrives at the aerial of the receiver via paths Pl to Px is filtered in a band-pass filter 4 in order to suppress signals outside the employed frequency band. The output signal of the band-pass filter 4 is amplified in an amplifier 5 and mixed in a mixer 6, at whose other input a local oscillator 7 is connected, to an inter-mediate frequency fIF This intermediate frequency signal now passes into the I/Q processor 2 where it is filtered in a further band-pass filter 8 and then mixed into the base band by multiplication (mixer 9, 9') by a cosine or sine signal of the frequency fo from a local oscillator 10. As a result, two resulting signals I(t) and Qtt) arise which are freed from high-frequency signal portions by low-pass filters 11, 11' and amplified in amplifiers 12, 12'. Subsequen~ly, the two signals are quantized in analog-to~digital converters 13, 13' with a sampling rate of fs = c/Tc and represented as digital values I(k) and Q(k) by b bits, b being for example equal to 8.
In the formula for the sampling rate fs, c desig-nates the number of sampling values per code element (=chip) and Tc the duration of a chip. This duration corresponds to the smallest square-wave pulse length of the auxiliary function used for band spreading, the characteristic of ~0 which of course is known to the receiver in a spread-spec-trum system and which is formed in the receiver by the reference code.
The third block of the receiver, the digital stage 3 which processes exclusively digital signals, consists ; essentially of correlators TIC, of a reference code gener-ator 14 and of a digital signal processor 15. The sequen-ces I(k) and Q(k) are each fed into a correlator TIC and correlated ~here with the reference code R(k).
ence code R(k). The correlation results CI~m) and CQ(m) are read by the digital signal processor 15 and processed further. The latter derives control signals PD from them for controlling the reference code generator 14 and the correlators TIC, calculates the deviation between the carrier frequency fIFand the local oscillator frequency fo and carries out the coherent demodulation of the transferred information bits. The reference code genera-tor 14 supplies the correlators TIC with the reference code R(k), the sampling rate fs and a synchronization signal SY, and supplies the analog-to-digital converters '' ' ' ' '' ' '., ' . .
_ 4 _ 2~
13, 13' with the sampling rake fs.
The requir~ments on digital correlators are det~rmined by the parameters of the in-house radio transmission system. A clock rate (chip rate) for the pseudo-random auxiliary function in the region of 10 to 30 ~z can ba derived from the coherence bandwidth of the propagation channel. In order to permit a sufficiently large number of users in a cellular system with multiple access (CDMA)spreading factors of at least 255 to approxi-mately 4000 are required. Since the synchronization of the local reference code to the received code should also occur digitally in a digital correlation receiver and in this procass each chip must be samplsd at least twice, sampling and processing rates for the correlator of 20 to 60 MHz are obtained.
Since a plurality of signals are superimposed on one another at the reception site due to multipath propa-gation and multiple occupation of the channel by means of CDMA, at the receiver input the signal no longer has a constant signal envelope and must therefore be amplitude-quantized in front of the correlator with resolution of sQ~7eral bits. If this does not occur, intolerable losses ~hen arise. The reference code can, however, be present as a binary signal.
A digital correlator which fulfils these require-ments can be realized as programmable transverse filter or as time-integrating correlator. ~n implementation as programmable transverse filter is known from the publica-tion ~Digital SOS-MOS Correlator: Basic System Component in Experimental Army Spread Spectrum Radio" by N.A. Saethe -cn, B. Skeie and S. Prytz in 2nd Int. Conf.
on the Impact of High Speed and VLSI Technology on Comm.
Systems, London 1983. However! his implementation is extremely costly since with two sampling values per code chip per signal bit at least 2.L memory cells for refer-ence code and signal and 2.L multipliers are xequired (L=code length). The dat~ rate of the correla-tion result is the same as the sampling rate of the input signal and therefore very high. This permits a rapid ~ 5 ~ 2~ 6 synchronization but is unfavourable for the further digital signal processing.
In contrast to the programmable transversal filter, the output data rate in a tlme-integrating correlator is reduced in relation to the sampling rate ~t the input by the number of sl ?A correlation products. With integration over a complete code period, the correlation results only occur - with the information bit rate, which signifies a reduction in the sampling rate of the input signal by the product c.L. (c=number of sampling values per code chip). This reduced data rate can now be easily processed further in a digital signal processor.
From the literature, implementations of time-integrating correlators in CCD technology (CCD: charge-coupled device) are known (B.E. Bur~e, D . L . Smythe: "ACCD Time-Integrating Correla~or", IEEE J. of Solid State Circuits, SC-18, Dec. 1983) and as acousto-optical com-ponents (F.B. Rotz: ~TLme-Integrating Optical Correla-tor~, Proc SPIE, Vol. 202, 1979). Since, on the one hand, the CCD solution is limited to clock rates of a -x;
of 20 MHz and has a dynamic range restricted by offset voltages and clock crosstalk, and, on the other hand, the acousto-optical solution cannot be built up monolithi-cally and is moreover complicated and expensive, these known implementations are not suitable for the digital multipath receiver of Fig. 1.
Although the digi~al correlator TIC is a time-integrating correlator, it is neither a charge-coupled nor an acousto-optical component, but rather its archi-tecture is especially adapted to the requirements of amultipath receiver according to the scanner principle.
The solution found for this which is described below is monolithically integrateable and therefore sLmple to use and it is space-saving and economical in terms of power. The programmable functions of the tLme-integrating correlator TIC and the evaluation of the correlation results in the digital signal processor 15 permit the construction of a very flexible spread~spectrum receiver.
Fig. 2 shows a block diagram of the time-. . .
. , ': .
.
~ .
_ 6 - ~ 6 integrating correlators TIC employed in the rec~iver o~
Fig. 1. The number sequences, designated in Fig. 1 by I'k) and Q(k), of the I/Q processor 2 are designated here in general as representative of both number sequences with S(k). According to the illustration, the correlator consists of N correlator stages Kn, of a control logic 16 and of a digital delay line 17, to which the correlator stages are connected in parallel.
The architectur~ of the time-integrating cor-relator is adapted to the expected impulse response of the propagation channel. This ir~ulse response is, as measure-ments have shown, significantly shorter than the data bit length, so that significant correlation values only arise during a short part of the code length. In the synchroni-~ed state, it is therefore sufficient to correlate only a small section of the complete code with the received signal. For this purpose, the binary reference code R(k) supplied by the reference code generator 14 (Fig. 1) is fed into the digital delay line 17, which has a certain length D, in the present case D=32, and which is operated with the clock fs, for example fs=2/Tc. The reference code is delayed hy the time Td between each of the t~rmin~ls of two successive correlator stages Kn, in which case preferably Td=Tc/2. The n-th correlator stage therefore receives the reference code delayed by n.Td as a re~erence signal.The di~itized output signal S(k) of the I/Q processor 2 (Fig. 1) having, for example, 8-bit resolution is connected in parallel to all corre-lator stages Kn.
Each correlator sta~e ~n contains, as illustra-ted, an accumul~tor 18 and a result memory 19 which are both controlled by the control logic 16, and calcu-lates the product of S(k) and R(k-n). These products are sl ~~ and stored in the respective accumulutor 18 35 over a complete code length k (k=lc times L). At the end of the summation, the correlation value Cn(m) is transmitted into the result memory 19 by a control signal RS' of the control logic 16, and then the accurnula~or 18 is set to zero again with RS. Each result memory 19 _ 7 _ 2~
supplies its correlation value Cn(m) after a correspond-ing control signal REn of the control logic 16 to the digital signal processor 15 (Fig. 1~ which further pro-cesses the results of the preceding correlation during the next summa~ian period. On the basis of the signal REn (Read Enable) and under the control of the digital signal processor in each case a single correlation is read out via a bus.
The design of the time-integrating correlator TIC
can be further simplified by means of a selection circuit 20 which connects a delay~d reference sequence for each correlation stage Kn with a delay which can be programmed by the digital signal processor lS. This simplification is based on the following consideration: the length D of the delay line 17 must be adapted to the ~ximll~ expected length of the channel impulse response which is approxim-ately 1 microsecond in the present case. However, in practice this impulse re~onse always consists of a plural-ity of discrete signal portions with specific delays tp, so that only these discrete signal proportions, there-fore, need to be correlated. For this purpose, correlator stages are placed only at those points tp where signal portions are effectively present. As a result, the number N of correlator stages can be kept substantially smaller than the length D of the delay line 17. In the present exemplary embodiment where D=32, a number of N=8 cor-relator stages is still sufficient. A precondition for this mode of operation is a free progralmmability of the delay of each correlator stage. The programming occurs through the digltal signal processor 15 which drives a line DL via control logic 16, -that programs the ~lection circuit 20 which, itself, conn~cts the reference sequence with the programmed delay to the resp~ctive correlator stage.
In addition to the already mentioned ad~antages of the digital multipath receiver, a further essential advantage consists in the f~ct that the essential func tions of a spread-spectrum receiver can be completely carried out by corresponding operations in the digital - 8 - ~0~ 6 signal processor. The receiver can be characterized by the following states:
- coarse synchronization (acquisition) - operation - re~synchronization on loss of the code synchronism.
As already mentioned, the duration of the irnpulse response the transmission channel is only a frhction of the data bit length and thus of the code length L. For the acquisition, the N correlator stages of the tLme-integrating correlator are now programmed in such a way that the delav positions be-tween succeeding stages differ by a constant delay e.Tc, e = 1 being selec-ted for example. By shifting the receive-side reference code by N.e.Tc after every accumulation ~eriod, t'.1~ complete code length is correlated sequentially with the received signal. The acquisition time Tacq is then Tacq=L.Tc.(L/N.e) in com-parison ~o L.Tc L for a correlator of the type of a pro-grammable transverse filterO The respective reception energy E~m) for each delay can be calculated from the correlation values CI(m), CQ(m) obtained in this way:
E(m)=CItm)2+CQ(m) 2 After searching the complete code length, the position is determined where the m~i reception energy has occurred. In order to confirm whether the energy -~i has really been found, it is e~ ;ned whether the energy around this -x; is grea~er by a specific amount than the noise energy averaged over the complete code length. If this is the case, the reference code generator 14 (Fis. 1) is programmed in such a way that the energy -Yi lies in the centre of the reception window covered by the correlator. Thus, the acquisition is ter~in~ted and the receiver goes into the normal operating state; otherwise the acquisition is repeated.
The normal operating state comprises the follow-ing functions: monitoring the channel ~ulse response (scanning); tracking the phase of the local reference code generator; estimating the carrier phase and coherent or differential demodulation of the signals of the individual reception paths; deriving the weighting _ 9 _ Z0~ i6 functions for the individual reception paths from the channel lmpulse response combining the individual paths and detecting the transferred data bit. If an error protection coding has occurred in the transmitter, the receiver can additionally supply a quality criterion for the following error decoder for ~he so-called soft decision.
The advantages of a multipath receiver then have the m~i effect if a good estimation of the channel impulse response can be carried out. This occurs in the present digital receiver by means of the scann; ng algo-rithm, for which S of ~he total N correla~or stages of the time-integrating corrPlator TIC can be used. The other correlator stages (number = N-S) are required in parallel to the latter for the data demodulation. With the scanner-correlation stages mutually ~nifted by Td, L.Tc correlation values are calculated during a code period. Then, the correlator stages are shifted by S.Td, and ~he calculation of the correlation values is repeated for this position. After ts=D/S code periods, the entire window is searched and the scanning process begins again. This scanning function is illustra-ted in Fig. 3a.
As can be seen in Fig. 3b, the correlator stages required for the demodulation are programmed at the points where the greatest reception power is to be axpected. These correlation values CI(m) and CQ(m) can be interpreted ~ccording to Fig.4a as coordinates of a data vector d(m) = (CI(m), CQ(m)) in the complex plane. Fig.
4a therefore constitutes the absolute value of the channel impulse response measured during the scanning ~unction.
That the measurements of the channel impulse response are significant when scAnning, the scan time ts must be smaller than the i n i change time of the transmission channel. ~he more correlator stages are used for the scAnning, the quicker the changing channels which can be monitored. Hence, correspondingly fewer correlator - ' . , .
stages are then available for the data demodulation. For typical in-house channels in the described system for example S=4 scanner channels are required. Since the time-integrating correlator TIC can be programmed by the digital signal processor 15 (Fig. 1), the partition into scanner and demodulator channels can also occur adaptively.
With the aid o~ the correlation values of the scanner channels, it is monitored whether the correlation 10 window of the time-integrating correlator TIC is cor-rectly positioned in relation to the received signal.
Because the channelimpulse response is usually shorter than the window width, a nois~ power can be detç ;ned from the values outside the impulse response. If the ratio 15 of the power of all the reception paths to this noise power falls below a predete ;ned threshold, a re-synchronization is initiated.
The fine synchronization (tracking~ of the reference code can occur in such a way that the weighted 20 average of the D correlation values, that is to say their median point, is positioned in the centre of the window of the time-integrating correlator. A different method consists in positioning the strongest reception path in each case at a specific point, for example at D.Td/3.
As has already been mentioned, the correlation values CI(m) and CQ(m) of the scanner channels can be interpreted coordinates of a data vector d(m) = (CI~m), CQ(m)) in the complex plane (Fig. 4a). The angle Phi between the data vector and the real axis then corres-30 ponds to the phase shift between the carrier of the received signal and the local oscillator. If the frequencies of the received signal and of the local oscillator coincide (fIF=fo), the angle Phi will remain constant on average and as~ume an average value and only 35 fluctuate around this average value due to the noise in the received signal.
For an optimum detection, by averaging over a plurality of reception vectors, the unit vector e=(xl, yl) is calculated for which the following applies:
.
.~
:
yl/xl=tan(phi) Z~ 6G
xl2+ylz=l Each received data vec~or d(m) is multiplied by the complex conjuga~e vector e:
S B(m)-Re(e .d(m)) From the real part of the product, the transmit-ted data bit is deterrin~d from the sign of B(m); the magnitude of the product is a measure for the reliability of the decision and can be supplied as quality informa-tion to a following error decoder.
If the frequencies of received signal and localoscillator are different, successive vectors d(m) and d(m+l) are rotated towards one another (Fig. 4b)o This rotation Psi is proportional to the frequency difference df and to the bit length Tb:
Psi=2.Pi.df.Tb In this case, in addition to the initial phase Phi, the frequency offset must also be estimated by means of the phase rot~tion Psi. From this, a re~erence vector r can be calculated which now no longer possesses a con-stant phase Phi, but rather rotates with the angular speed 2.Pi.df:
r(m)=(x2, y2) y2/x2=tan(Phi~2.PiOdf.m.Tb)~5 The demodulation occurs in the same way as before:
B(m)-Re(r(m) .d(m)) The estimation of the frequency difference df can occur, for example, by means of a Fast Fourier transform (FFT) by means of 2M successive data vectors d(m).
(m=1... 2M, to example m=1... 32).
In a multipath receiver, each reception path n usually has a different phase shift Phi in relation to the local oscilla~or because of the different lengths of the propagation paths. The frequenc~ difference df is, however~ almost identical in in-house channels fvr all paths, since Doppler effect can be neglected. Therefore, the estimation of the frequency difference for all paths can occur jointly, whilst the phase for each path must be calculated individually. A simple solution is - 12 - 20~
offered here by the differential demodulation of two successive data vec~ors d(m)-1) and d(m). Here, the first vector d(m-1) is rotated by the amount Psi=2.Pi.df.Tb and then used as reference for ~he demodula~ion of the next data vector d(m):
r(m)=d(m-l).e The demodulated bit value is:
B(m)=Re(r(m) .d(m)) The demodulated bit values B(m) of n channels are present in the multipath receiver. Said values have to be combined in an appropriate manner in order to enable a decision relating to the received bit to be made. For this purpose, for each reception channel the expected value of the signal amplitude Gn=[(dn(m)-dn(m)*)~] is calculated as weighting. The bit values B(m) are multi~
plied by the square of Gn and the produc~s are summed over all values of n to form S(m). ~he sign of S(m) yi~lds the value of the detected bit and the magnitude of S(m) is a measure of the strength of the received signal and therefore also of the reliability of this decision.
If the code synchronism is lost in operation, a resynchronization is performed. In contrast to the acguisition, however, the complete code length is not searched, but rather one of the search algorithms known from the literature is used, for example that according to the publication ~Performance Analysis for the Expand-ing Search PN Acquisition Algorithm" by W.R. Braun, IEEE
Trans. Comm., COM-30, 1982.
Thus, all the essential functions of the mul~i-path receiver, in particular the evaluation of theresults of the correlation have been carried out in the digital signal processor, which, in con~unction with the programmability of the functions of the time-integrating correlator, gives the described multipath receiver a considerable flexibility. In addition, the use o~ the time-integrating correlator with the described architec-ture permits a monolithically integrateable solution which is easy to use, space-saving and economical in terms of power.
Claims (17)
1. Digital receiver for direct-sequence-spread-spectrum signals generated by multiplication of an information-carrying signal by an auxiliary function, having a correlator for the correlation of the received signal with a reference code, and having a digital signal processor for evaluating the results of the correlation, characterized by a digital, time-integrating, multistage correlator (TIC), the individual stages of which are connected on the one hand to a digital delay line (17) fed with the reference code (R(k)), and to which on the other hand a digital signal (S(k)) recovered from the received signal is applied, the length (D) of the delay line being adapted to the maximum expected length of the impulse response of the transmission channel.
2. Receiver according to Claim 1, characterized in that in each correlator stage (Kn) the product of the aforesaid digital signal (S(k)) and the corresponding delayed reference code (R(k-n)) is calculated, summed and stored over a code length.
3. Receiver according to Claim 2, characterized in that the estimation of the channel impulse response is performed by a scanning algorithm for which a programmable number (S) of the total number (N) of the correlator stages (Kn) is used, the correlator stages used for the scanning being shifted by a delay (Td) respectively.
4. Receiver according to Claim 3, characterized in that during a code period of codelength times duration of a code element correlation values are calculated with the correlator stages used for the scanning, that afterwards the correlator stages are shifted by an amount of number (S) of the correlator stages used for the scanning algorithm times the delay (Td), that the correlation values for this new position are calculated, and that the said shift occurs until the entire width of the scanning window is covered, whereupon the algorithm is started again.
5. Receiver according to Claim 4, characterized in that the correlator stages used for the demodulation are positioned on places with the highest absolute values of the channel impulse response found by the scanning algorithm.
6. Receiver according to Claim 5, characterized in that the partition of the total number (N) of the correlator stages (Kn) into a number (S) for the scanning and a number (N-S) for the demodulations is performed dynamically, depending on changing characteristics of the propagation channel.
7. Receiver according to any of Claims 2 to 6, characterized in that each correlator stage (K(n)) has an accumulator (18) and a result memory (19), the latter being provided for the controlled output of the correlation value stored in it to the digital signal processor (15).
8. Receiver according to Claim 7, characterized in that the number (N) of the correlator stage (Kn) corresponds to the length (D) of the delay line (17).
9. Receiver according to Claim 7, characterized in that the number (N) of correlator stages (Kn) is smaller than the length (D) of the delay line (17), and in that the correlator stages are arranged only at the points of the occurrence of discrete signal portions of the impulse response, the delay (Td) of each correlator stage being freely programmable.
10. Receiver according to Claim 9, characterized by a selection circuit (20), provided between the delay line (17) and the correlator stages (Kn), for connection the reference sequence (R(k)) to the respective correlator stage with the corresponding programmed delay.
11. Receiver according to Claim 8 or 9, characterized in that the delay line (17) is operated with a clock corresponding to the sampling rate (fs) during the digitization of the received signal.
12. Receiver according to Claim 11, characterized by a converter (1) for mixing the filtered received signal to an intermediate frequency (fIF) and by a stage (2) connected downstream of the converter, in which stage the intermediate frequency signal is multiplied by a cosine signal and a sine signal from a local oscillator (10) and mixed into the base band, two resulting signals (I(t), Q(t)) arising.
13. Receiver according to Claim 12, characterized in that the aforesaid stage (2) has an analog-to-digital converter (13, 13') for each of the resulting signals (I(t), Q(t)).
14. Receiver according to Claim 11, characterized in that a sequential correlation of the received signal with the reference code (R(k)) occurs for the coarse synchronization, the position between the individual correlator stages (Kn) differing by a constant delay, and the local reference code being shifted after each correlation by an amount corresponding to the product of this constant delay times the number (N) of the correlator stages.
15. Receiver according to Claim 14, characterized in that a calculation of the respective reception energy for each delay occurs from the correlation values (CI(m), CQ(m)) received during the sequential correlation, and the energy maximum is determined, and in that the reference code generator (14) is controlled in such a way that this energy maximum lies in the centre of the reception window which can be covered by the time-integrating correlator (TIC).
16. Receiver according to Claim 15, characterized in that the correlation values (CI(m), CQ(m)) form the components of a reception vector (d(m)) in the complex plane (I/Q), and that a unit vector is formed from a plurality of successive reception vectors and each reception vector is multiplied by the complex conjugate of the unit vector, and in that the transmitted data bit is determined from the sign of this product.
17. Receiver according to Claim 16, characterized in that the unit vector is formed by averaging, and in that, before this averaging, a compensation of the phase rotation (Psi), brought about by frequency deviations between reception frequency and local oscillator frequency, of each reception vector (d(m)) occurs.
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| CH3811/89-0 | 1989-10-19 | ||
| CH3811/89A CH679718A5 (en) | 1989-10-19 | 1989-10-19 |
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| CA2044166C true CA2044166C (en) | 1998-02-03 |
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| CA002044166A Expired - Fee Related CA2044166C (en) | 1989-10-19 | 1990-10-05 | Digital receiver for spread-spectrum signals |
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| US (1) | US5216691A (en) |
| EP (1) | EP0448665B1 (en) |
| AT (1) | ATE122189T1 (en) |
| CA (1) | CA2044166C (en) |
| CH (1) | CH679718A5 (en) |
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| US10957445B2 (en) | 2017-10-05 | 2021-03-23 | Hill-Rom Services, Inc. | Caregiver and staff information system |
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4164628A (en) * | 1977-06-06 | 1979-08-14 | International Telephone And Telegraph Corporation | Processor for multiple, continuous, spread spectrum signals |
| US4361891A (en) * | 1980-12-22 | 1982-11-30 | General Electric Company | Spread spectrum signal estimator |
| US4530103A (en) * | 1983-08-22 | 1985-07-16 | E-Systems, Inc. | Method and apparatus for baseband tracking of a PN code sequence in a spread spectrum receiver |
| US4611333A (en) * | 1985-04-01 | 1986-09-09 | Motorola, Inc. | Apparatus for despreading a spread spectrum signal produced by a linear feedback shift register (LFSR) |
| US4755969A (en) * | 1986-11-07 | 1988-07-05 | Digital Electronic Communications Equipment (Dece Corp.) | Pseudo random sequence generation |
| IT1206820B (en) * | 1987-06-26 | 1989-05-03 | Selenia Spazio Spa | DIRECT SEQUENCE DIGITAL "SPREAD SPECTRUM" MODEM |
| US4813006A (en) * | 1987-06-29 | 1989-03-14 | Hughes Aircraft Company | Analog-digital correlator |
| CA1293999C (en) * | 1987-08-24 | 1992-01-07 | Osamu Ichiyoshi | Earth station capable of effectively using a frequency band of asatellite |
| US4894842A (en) * | 1987-10-15 | 1990-01-16 | The Charles Stark Draper Laboratory, Inc. | Precorrelation digital spread spectrum receiver |
| DE3735374A1 (en) * | 1987-10-19 | 1989-05-03 | Siemens Ag | Digital correlator circuit |
| CH676179A5 (en) * | 1988-09-29 | 1990-12-14 | Ascom Zelcom Ag | |
| JPH0738089B2 (en) * | 1988-11-30 | 1995-04-26 | 三田工業株式会社 | Developer container |
| GB8916623D0 (en) * | 1989-07-20 | 1989-09-06 | Ncr Co | Spread spectrum signal demodulator |
| US5134407A (en) * | 1991-04-10 | 1992-07-28 | Ashtech Telesis, Inc. | Global positioning system receiver digital processing technique |
-
1989
- 1989-10-19 CH CH3811/89A patent/CH679718A5/de not_active IP Right Cessation
-
1990
- 1990-10-05 US US07/689,820 patent/US5216691A/en not_active Expired - Fee Related
- 1990-10-05 ES ES90914499T patent/ES2074170T3/en not_active Expired - Lifetime
- 1990-10-05 DK DK90914499.0T patent/DK0448665T3/en not_active Application Discontinuation
- 1990-10-05 EP EP90914499A patent/EP0448665B1/en not_active Expired - Lifetime
- 1990-10-05 DE DE59009021T patent/DE59009021D1/en not_active Expired - Fee Related
- 1990-10-05 CA CA002044166A patent/CA2044166C/en not_active Expired - Fee Related
- 1990-10-05 WO PCT/CH1990/000237 patent/WO1991006164A1/en active IP Right Grant
- 1990-10-05 AT AT90914499T patent/ATE122189T1/en not_active IP Right Cessation
Also Published As
| Publication number | Publication date |
|---|---|
| ATE122189T1 (en) | 1995-05-15 |
| EP0448665B1 (en) | 1995-05-03 |
| US5216691A (en) | 1993-06-01 |
| ES2074170T3 (en) | 1995-09-01 |
| CH679718A5 (en) | 1992-03-31 |
| DK0448665T3 (en) | 1995-09-25 |
| CA2044166A1 (en) | 1991-04-20 |
| DE59009021D1 (en) | 1995-06-08 |
| EP0448665A1 (en) | 1991-10-02 |
| WO1991006164A1 (en) | 1991-05-02 |
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| Date | Code | Title | Description |
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| EEER | Examination request | ||
| MKLA | Lapsed | ||
| MKLA | Lapsed |
Effective date: 20031006 |