US9270325B2 - Iterative interference suppression using mixed feedback weights and stabilizing step sizes - Google Patents
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
- H04B1/7097—Interference-related aspects
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
- H04B1/7097—Interference-related aspects
- H04B1/7103—Interference-related aspects the interference being multiple access interference
- H04B1/7107—Subtractive interference cancellation
- H04B1/71075—Parallel interference cancellation
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/69—Spread spectrum techniques
- H04B1/707—Spread spectrum techniques using direct sequence modulation
- H04B1/7097—Interference-related aspects
- H04B1/711—Interference-related aspects the interference being multi-path interference
- H04B1/7115—Constructive combining of multi-path signals, i.e. RAKE receivers
- H04B1/712—Weighting of fingers for combining, e.g. amplitude control or phase rotation using an inner loop
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B2201/00—Indexing scheme relating to details of transmission systems not covered by a single group of H04B3/00 - H04B13/00
- H04B2201/69—Orthogonal indexing scheme relating to spread spectrum techniques in general
- H04B2201/707—Orthogonal indexing scheme relating to spread spectrum techniques in general relating to direct sequence modulation
- H04B2201/70702—Intercell-related aspects
Definitions
- Patent Application Publication Number 2005-0180364 A1 which incorporates by reference and is a Continuation-in-Part of (a) U.S. patent application Ser. No. 11/773,777, entitled “Systems and Methods for Parallel Signal Cancellation,” and filed on Feb. 6, 2004, now U.S. Pat. No. 7,394,879; (b) U.S. patent application Ser. No. 10/686,359, entitled “Systems and Methods for Adjusting Phase,” and filed Oct. 15, 2003, now U.S. Pat. No. 7,068,706; (c) U.S. patent application Ser. No. 10/686,829, entitled “Method and Apparatus for Channel Amplitude Estimation and Interference Vector Construction,” and filed on Oct.
- a communication resource is divided into code-space subchannels that are allocated to different users.
- a plurality of sub channel signals received by a wireless terminal may correspond to different users and/or different sub channels allocated to a particular user.
- a single transmitter broadcasts different messages to different receivers, such as a base station in a wireless communication system broadcasting to a plurality of mobile terminals
- the channel resource is subdivided in order to distinguish between messages intended for each mobile.
- each mobile terminal by knowing its allocated subchannel(s), may decode messages intended for it from the superposition of received signals.
- a base station typically separates received signals into subchannels in order to differentiate between users.
- received signals are superpositions of time-delayed and complex-scaled versions of the transmitted signals. Multipath can cause several types of interference. Intra-channel interference occurs when the multipath time-delays cause subchannels to leak into other subchannels.
- subchannels that are orthogonal at the transmitter may not be orthogonal at the receiver.
- inter-channel interference caused by unwanted signals received from other base stations.
- Each of these types of interference can degrade communications by causing a receiver to incorrectly decode received transmissions, thus increasing a receiver's error floor.
- Interference may also have other deleterious effects on communications. For example, interference may lower capacity in a communication system, decrease the region of coverage, and/or decrease maximum data rates. For these reasons, a reduction in interference can improve reception of selected signals while addressing the aforementioned limitations due to interference.
- the wireless transmission medium is characterized by a time-varying multipath profile that causes multiple time-delayed replicas of the transmitted waveform to be received, each replica having a distinct amplitude and phase due to path loss, absorption, and other propagation effects.
- the received code set is no longer orthogonal.
- the code space suffers from intra-channel interference within a base station as well as inter-channel interference arising from transmissions in adjacent cells.
- the Rake receiver uses a channel-tracking algorithm to resolve the received signal energy onto various multipath delays. These delayed signals are then weighted by the associated complex channel gains (which may be normalized by path noise powers) and summed to form a single resolved signal, which exploits some of the path diversity available from the multipath channel. It is well known that the Rake receiver suffers from a significant interference floor, which is due to both self-interference from the base station of interest (or base stations, when the mobile is in a soft-handoff base station diversity mode) and multiple-access interference from all base stations in the coverage area. This interference limits the maximum data rates achievable by the mobiles within a cell and the number of mobiles that can be supported in the cell.
- the optimal multi-user detector has the best performance, but is generally too computationally complex to implement. MUD complexity increases exponentially with respect to the total number of active sub channels across the cell of interest and the interfering cells as well as the constellation size(s) of the subchannels. This complexity is so prohibitive that even efficient implementations based on the Viterbi algorithm cannot make it manageable in current hardware structures.
- Another approach is a properly designed linear receiver, which in many channel scenarios, is able to retain much of the optimal MUD performance, but with a complexity that is polynomial in the number of subchannels.
- LMMSE linear minimum mean squared error
- decorrelating or zero-forcing
- PNA-LMMSE PN-averaged LMMSE
- This receiver is generally inferior to the LMMSE approach, it has the advantage of not having to be implemented directly, because it is amenable to adaptive (or partially adaptive) implementations.
- the advantages of an adaptive implementation over a direct implementation include reduced complexity and the fact that the additive noise power (i.e., background RF radiation specific to the link environment, noise in the receiver's RF front end, and any processing noise such as noise due to quantization and imperfect filtering) does not have to be estimated.
- PIC parallel interference cancellation
- the interference-cancelled subchannels are then fed to a subsequent PIC stage. Ideally, within just a few stages (i.e., before the complexity grows too large), the performance rivals that of the full linear receiver using a matrix inverse.
- PIC can be implemented in various modes depending on what types of symbol estimates are used for interference cancellation.
- a soft-cancellation mode PIC does not exploit additional information inherent in the finite size of user constellations. That is, estimates of data symbols are not quantized to a constellation point when constructing interference signals.
- the user constellations maybe known (e.g., in an EV-DO link or in a WCDMA link without HSDPA users) or determined through a modulation classifier. In such cases, it is possible for PIC to be implemented in a hard-cancellation mode. That is, estimates of data symbols are quantized to constellation points (i.e., hard decisions) when constructing the interference signal.
- PIC In a mixed-cancellation mode, PIC employs a soft decision on each symbol whose constellation is unknown, and either a soft or hard decision on each symbol whose constellation is known, depending on how close the soft estimate is to the hard decision.
- Such a mixed-decision PIC typically outperforms both the soft-decision PIC and the hard-decision PIC. Moreover, it can also substantially outperform the optimal LMMSE receiver and promises even greater performance gains over PNA-LMMSE approaches currently under development for advanced receivers. The performance of soft-decision PIC is bounded by the optimal LMMSE.
- embodiments of the present invention may provide a generalized interference-canceling receiver for canceling intra-channel and inter-channel interference in coded, multiple-access, spread-spectrum transmissions that propagate through frequency-selective communication channels.
- Receiver embodiments may employ a designed and/or adapted soft-weighting subtractive cancellation with a stabilizing step-size and a mixed-decision symbol estimator.
- Receiver embodiments may be designed, adapted, and implemented explicitly in software or programmed hardware, or implicitly in standard Rake-based hardware, either within the Rake (i.e., at the finger level) or outside the Rake (i.e., at the sub channel symbol level).
- Embodiments of the invention may be employed in user equipment on the forward link and/or in a base station on the reverse link.
- Some embodiments of the invention address the complexity of the LMMSE approach by using a low-complexity iterative algorithm. Some embodiments of the invention in soft-mode may be configured to achieve LMMSE performance (as contrasted to the lesser-performing PNA-LMMSE) using only quantities that are easily measured at the receiver. Some embodiments address the sub-optimality of the LMMSE and PNA-LMMSE approaches by using an appropriately designed mixed-decision mode and may even approach the performance of an optimal multi-user detector. In some embodiments, stabilizing step sizes may be used to enhance stability of various PIC approaches. Some embodiments may employ symbol-estimate weighting to control convergence of various PIC approaches.
- Some embodiments of the invention address the limitation of various PIC approaches to binary and quaternary phase shift keying in mixed-decision mode by being configurable to any sub channel constellation. Some embodiments of the invention address the difficulty of efficiently implementing various PIC approaches in hardware by using a modified Rake architecture. Some embodiments of the invention address the so-called “ping-pong effect” (i.e., when the symbol error rate oscillates with iteration) in various PIC approaches by pre-processing with a de-biasing operation when making symbol estimates.
- the weight-calculation means may include, by way of example, but without limitation, any combination of hardware and software configured to calculate symbol weights from a function employing a merit of at least one input symbol decision.
- the merit may comprise an average ratio of signal power to interference-plus-noise power (or a function thereof).
- the merit may be a function of input symbol decisions and proximity of those input symbol decisions to a nearby constellation point.
- the weight-calculation means may employ time-series averaging for calculating the proximity as a statistical average.
- the weight-calculation means may include a signal processing means configured to perform statistical signal processing for estimating the average ratio of signal power to interference-plus-noise power. Such statistical signal processing may employ error-vector magnitude calculations.
- the synthesizing means is configured for employing a signal basis for all symbol sources in the channel to synthesize constituent signals from the plurality of weighted symbol estimates.
- the synthesizing means may combine the constituent signals for producing a combined signal.
- the combined signal may be considered as approximating a received signal that would result from signals transmitted by at least one hypothetical transmitter configured to transmit signals synthesized from the plurality of weighted symbol estimates.
- the synthesizing means may include, by way of example, but without limitation, a signal processor or a set of elements of a transmitter configured to process symbol estimates for producing at least one modulated digital baseband signal, such as a signal that may be produced by a transmitter prior to up-conversion, amplification, and coupling into a communication channel.
- the synthesizing means comprises a Walsh coder (such as a fast Walsh transform) and a pseudo-noise (PN) coder.
- the subtraction means is configured to subtract the combined signal from the received signal to produce an error signal.
- the subtraction means may include, by way of example, but without limitation, an adder, a combiner, or any other device or method configured for subtracting a first set of signals from a second set of signals.
- the stabilizing step size means is configured for scaling the error signal by a stabilizing step size to produce a scaled error signal.
- the stabilizing step size means may include, by way of example, but without limitation, any combination of hardware and software configured to scale an error signal with a scaling factor that may be used for controlling convergence in an iterative canceller.
- the stabilizing step-size means may comprise a step size calculation means and a multiplier means for scaling an error signal with the step size.
- the step size calculation means is configured for calculating a stabilizing step size having a magnitude that is a function of proximity of the input symbol decisions to a desired interference-cancelled symbol decision.
- the multiplier means is configured for scaling (e.g., multiplying) an error signal with the stabilizing step size.
- the step size calculation means may include, by way of example, but without limitation, software or programmable hardware configured for calculating a stabilizing step size.
- the resolving means is configured for resolving a scaled error signal or each of a plurality of interference-cancelled constituent signals onto a signal basis for all of symbol sources in a channel.
- the resolving means may include, by way of example, but without limitation, a channel estimator, a PN decoder, and a Walsh decoder.
- 3G TS 25.211, 3G TS 25.212, 3G TS 25.213, and 3G TS 25.214 (the WCDMA standard), (4) the standard offered by a consortium named “3rd Generation Partnership Project 2” (3GPP2) and embodied in a set of documents including “TR-45.5 Physical Layer Standard for cdma2000 Spread Spectrum Systems,” the “C.S0005-A Upper Layer (Layer 3) Signaling Standard for cdma2000 Spread Spectrum Systems,” and the “C.S0024 CDMA2000 High Rate Packet Data Air Interface Specification” (the CDMA2000 standard), (5) Multi-Code CDMA systems, such as High-Speed-Downlink-Packet-Access (HSDPA), and (6) other CDMA standards.
- 3GPP2 3rd Generation Partnership Project 2
- FIG. 2 is a block diagram illustrating a front-end processor for an iterative interference canceller.
- FIG. 8A is a block diagram illustrating post interference-cancellation signal despreading on constituent finger signals.
- FIG. 1 Various functional elements or steps, separately or in combination, depicted in the figures may take the form of a microprocessor, digital signal processor, application specific integrated circuit, field programmable gate array, or other logic circuitry programmed or otherwise configured to operate as described herein. Accordingly, embodiments may take the form of programmable features executed by a common processor or discrete hardware unit.
- the following formula represents an analog baseband signal received at a mobile from multiple base stations, each with its own multipath channel,
- FIG. 1 is a block diagram of an iterative interference canceller (IIC), which is a low-complexity receiver configured to mitigate intra-channel and inter-channel interference.
- the received baseband signal y(t) is input to a front-end processor 101 , which produces initial symbol estimates for all symbols of the active users served by at least one base station.
- the initial symbol estimates are coupled to a first interference cancellation unit (ICU) 102 configured to cancel a portion of the intra-channel and inter-channel interference that corrupts the symbol estimates.
- the ICU 102 outputs a first set of updated symbol estimates, which are interference-cancelled symbol estimates.
- the updated symbol estimates are coupled to a second ICU 103 .
- a plurality M of ICUs 102 - 104 illustrate an iterative process for performing interference cancellation in which the initial symbol estimates are updated M times.
- slice ( ⁇ circumflex over (b) ⁇ (s), k [i] ) represents the symbol estimate ⁇ circumflex over (b) ⁇ (s), k [i] sliced (i.e., quantized) to the nearest constellation point from which the symbol b (s), k was drawn.
- This approach is applicable for symbols with known constellations. For example, it is typical for a receiver to know the symbol constellation for a user of interest, but it may not know which constellations are assigned to other users.
- Equation ⁇ ⁇ 8 The subscript k on the left-hand side denotes that the constituent signal is for a user k, whereas the subscript l on the left-hand side of Equation 6 represents that the constituent signal is for a finger l.
- the sum of the user constituent signals produces a synthesized received signal
- Equation ⁇ ⁇ 9 The left-hand sides of Equation 7 and Equation 9 are the same signal, whereas the right-hand sides are simply two different decompositions.
- FIG. 6A shows a cancellation module 601 (such as the canceller 303 in FIG. 3 ) configured to perform interference cancellation 610 on constituent signals, followed by Rake processing and despreading 611 in a Rake-based receiver.
- FIG. 6B shows a cancellation module 602 configured to synthesize 621 a received signal from constituent components, followed by Rake processing and despreading 622 , and interference cancellation 623 .
- FIG. 7 is a block diagram of an interference canceller comprising a plurality B of cancellers 701 - 703 configured to perform interference cancellation on a plurality J of constituent signals for each of a plurality B of base stations. Since the constituents signals may be either fingers or users, index j ⁇ 0, 1, . . . , J (s) ⁇ 1 ⁇ is expressed by
- the scaled residual signal is combined with the constituent signals ⁇ tilde over (y) ⁇ (s), j [i] in combining modules 712 - 714 to produce a set of interference-cancelled constituents represented by z (s),j [i] ( t ) ⁇ ⁇ tilde over (y) ⁇ (s),j [i] ( t )+ ⁇ [i] ( y ( t ) ⁇ ⁇ tilde over (y) ⁇ [i] ( t )), Equation 10 where z (s), j [i] (t) is an interference-cancelled j th constituent signal for base stations (s).
- J (s) ⁇ ⁇ 0, 1, . . . , J (s) ⁇ 1 ⁇ are indices of the subset of constituent signals to be employed in cancellation.
- Embodiments of the invention may be configured for applications in which hardware limitations restrict the number of finger signals or user signals that can be used for interference cancellation (e.g., only the strongest constituents are used).
- Interference cancelled signals z (s), l [i] (t) are time-advanced 810 - 811 by an amount ⁇ (s), l .
- a maximal ratio combining module scales 812 - 813 each time-advanced a signal z (s), l [i] (t+ ⁇ (s), l ) by ⁇ (s), l */ ⁇ ⁇ (s) ⁇ and combines 814 the time-advanced signals for each base station.
- a resolving module comprising multipliers 815 - 816 and integrators 817 - 818 resolves each combined signal
- Interference cancelled signals z (s), k [i] (t) corresponding to a k th user and s th base station are processed by a plurality L (s) of time-advance modules 831 - 832 corresponding to the multipath channel for the s th base station.
- a resolving module comprising multiplier 836 and integrator 837 resolves the combined signal onto the k th user's code waveform to give
- Equation ⁇ ⁇ 12 The values ⁇ tilde over (b) ⁇ (s), k [i] shown in Equation 11 and Equation 12 are generally not the same value, since the value of ⁇ tilde over (b) ⁇ (s), k [i] in Equation 11 is produced by cancellation employing finger constituents, whereas ⁇ tilde over (b) ⁇ (s), k [i] expressed by Equation 12 is produced by cancellation employing user constituents.
- FIG. 9A is a block diagram of a Rake despreader, such as Rake despreaders 611 and 622 shown in FIGS. 6A and 6B , respectively.
- the Rake despreader comprises a plurality B of Rake despreading modules 901 - 903 , each configured to process constituent signals from one of a plurality B of base stations.
- An exemplary Rake despreader module 902 is a block diagram illustrating functionality of each of the Rake despreader modules 901 - 903 .
- Input constituent signals ⁇ tilde over (y) ⁇ (s), j [i] (t) for all values of j are subtracted 911 from the received signal y(t) to produce a difference signal, or error signal, representing the difference between the received signal and the synthesized estimates of signals received by the base stations.
- the difference signal is y(t) ⁇ tilde over (y) ⁇ (s) [i] (t), where
- the difference signal for base station (s) is processed by a parallel bank of time advance modules 912 - 913 associated with the multipath channel for that base station, followed by maximal-ratio combining.
- a maximal-ratio combining module is configured to perform weighting 914 - 915 and combining 916 .
- a resolving module comprising multipliers 917 - 918 and integrators 919 - 920 resolves the resulting combined signals onto code waveforms of the base station's users to give the difference signal vector, or error signal vector, q (s), k ⁇ tilde over (q) ⁇ (s), k [i] , where
- Rake despreading such as described with respect to the exemplary Rake despreading module 902 , may also be accomplished explicitly by employing matrix multiplication to synthesize constituent signals of the received signal, such as represented by block 931 shown in FIG. 9B .
- a diagonal soft-weighting matrix may be defined as
- ⁇ [ i ] diag ( ⁇ ( 0 ) , 0 [ i ] , ... ⁇ , ⁇ ( 0 ) , K ( 0 ) - 1 [ i ] ⁇ ⁇ ⁇ ... ⁇ ⁇ ⁇ ⁇ ( B - 1 ) , 0 [ i ] , ... ⁇ , ⁇ ( B - 1 ) , K ( B - 1 ) - 1 [ i ] ) Equation ⁇ ⁇ 14 in which all of the users' soft weights are ordered first by base station and then by users within a base station. The same indexing may also be used to express the column vector of symbol estimates input to an i th ICU as
- Equation ⁇ ⁇ 15 The weighted symbol estimates are expressed as ⁇ [i] b [i] , and the outputs of the Rake despreading modules 901 - 903 are expressed by the difference equation,
- Equation ⁇ ⁇ 18 The values of q [i] represent the despread signals, such as described with respect to FIG. 2 .
- the values of ⁇ tilde over (q) ⁇ [i] are represented by Equation 13, and R is a square matrix whose elements are correlations between the users' received code waveforms.
- Equation 16 the functionality expressed by Equation 16 is implemented via the matrix-multiplication block 931 and a subtraction module 932 .
- Equation 14 B - 1 ⁇ ⁇ K ( s ) is employed for ordering users (first by base station, and then by users within a base station) described with respect to Equation 14.
- a user ⁇ is a member of base station (s) and a user ⁇ ′ is a member of base station (s′)
- the ( ⁇ , ⁇ ′) element of matrix R may be expressed by
- the elements of R can be built at the receiver with estimates of the path gains, path delays, and knowledge of the users' code waveforms.
- FIG. 10 is a block diagram of an interference canceller, such as the interference-cancellation block 623 shown in FIG. 6 .
- the difference signal q (s), k ⁇ tilde over (q) ⁇ (s), k [i] is scaled with a stabilizing step size ⁇ (i) by a stabilizing step size module 1001 , which may include a calculation module (not shown) configured to calculate a stabilizing step size having a magnitude that is a function of proximity of input symbol decisions to a desired interference-cancelled symbol decision.
- F is a block-diagonal matrix with a plurality B of diagonal blocks, wherein an s th diagonal block is a K (s) ⁇ K (s) block representing the users' transmit correlation matrix for base station (s).
- the stabilizing step size ⁇ [i] may be used to enhance interference cancellation in each ICU and/or stabilize iterative interference cancellation.
- a quality metric of a canceller's output ⁇ tilde over (b) ⁇ [i+1] may be derived as follows. If it is known (or approximated) that the additive noise w(t) in Equation 1 is Gaussian, then the despread outputs q , conditional on the transmitted symbols
- FIG. 11 A illustrates a method and apparatus for calculating a stabilizing step size.
- a Rake receiver 1100 comprises a first Rake, maximal ratio combining, and despreading unit 1101 to process a received signal y(t) for producing an output despread signal vector q .
- a second Rake, maximal ratio combiner, and despreader unit 1102 processes a synthesized receive signal with weighted symbol estimates corresponding to an i th iteration, and represented by
- a combiner 1103 calculates the difference between the outputs of 1101 and 1102 to produce a difference signal, or error signal, ⁇ [i] ⁇ q ⁇ R ⁇ [i] ⁇ circumflex over (b) ⁇ [i] , whose elements are indexed first by base station, and then by users within a base station,
- ⁇ _ [ i ] [ ⁇ ( 0 ) , 0 [ i ] , ... ⁇ , ⁇ ( 0 ) , K ( 0 ) - 1 [ i ] ⁇ ⁇ ⁇ ... ⁇ ⁇ ⁇ ⁇ ( B - 1 ) , 0 [ i ] , ... ⁇ , ⁇ ( B - 1 ) , K ( B - 1 ) - 1 [ i ] ] T .
- a difference signal y(t) ⁇ tilde over (y) ⁇ (s) [i] (t) may be produced prior to despreading, such as shown by block 1110 in FIG. 11B .
- Equation 24 becomes
- ⁇ [ i ] ( q _ - R ⁇ ⁇ ( ⁇ [ i ] ) 2 ⁇ b ⁇ _ [ i ] ) H ⁇ ⁇ [ i ] ⁇ ( q _ - R ⁇ ⁇ ⁇ [ i ] ⁇ b ⁇ _ [ i ] ) ( q _ - R ⁇ ⁇ ⁇ [ i ] ⁇ b ⁇ _ [ i ] ) H ⁇ ( ⁇ [ i ] ) H ⁇ R ⁇ ⁇ ⁇ [ i ] ⁇ ( q _ - R ⁇ ⁇ ⁇ [ i ] ⁇ b ⁇ _ [ i ] ) , Equation ⁇ ⁇ 26
- the signal ⁇ [i] is generated by a Rake, maximal ratio combining, and despreading unit 1120 and multiplied 1121 by ⁇ [i] to produce vector ⁇ [i] ⁇ [i]
- a synthesized received signal is generated 1124 from the vector ( ⁇ [i] ) 2 ⁇ [i] and processed with received signal y(t) by an adder 1125 to produce a difference signal.
- a Rake/combiner/despreader 1126 processes the difference signal to generate the vector q ⁇ R( ⁇ [i] ) 2 ⁇ circumflex over (b) ⁇ [i] .
- the inner product 1127 between this vector and the vector ⁇ [i] ⁇ [i] gives the numerator of Equation 26.
- the stabilizing step size may be derived from the multi path channel gains
- FIG. 12 is a block diagram of a symbol-estimation block comprising a plurality B of mixed-decision modules 1201 - 1203 configured to process signals received from B base stations.
- Mixed-decision module 1202 shows functionality that is common to all of the mixed-decision modules 1201 - 1203 .
- De-biasing modules 1210 - 1211 scale each of a plurality K (s) of input symbol estimates ⁇ tilde over (b) ⁇ (s), k [i+1] with non-negative de-biasing comtant d (s), k [i] for producing de-biased input symbol estimates.
- the mixed-decision module 1202 includes symbol-estimation modules 1212 - 1213 configured to perform symbol estimation on de-biased input symbol estimates whose constellations are known at the receiver.
- the de-biasing constant may be expressed by
- each value s (s), k [i] ⁇ tilde over (b) ⁇ (s), k [i+1] is operated on by a map ⁇ (s), k that takes the input into the complex plane to yield the updated symbol estimate
- the map ⁇ (s), k may be a mixed-decision map, which is a combination of soft and hard decisions.
- a soft-decision map is provided by a function ⁇ (s), k (x) that is a continuous function whose output ranges over the complex plane. Common examples, include, but are not limited to,
- some metric e.g., Euclidean distance
- a mixed-decision map ⁇ (s), k mixed (x) produces an output that is a soft decision or a hard decision, such as
- ⁇ ( s ) , k mixed ⁇ ( x ) ⁇ ⁇ ( s ) , k hard ⁇ ( x ) if ⁇ ⁇ SIN ⁇ R ( s ) , k > c ( s ) , k ⁇ ( s ) , k soft ⁇ ( x ) otherwise Equation ⁇ ⁇ 34
- the mixed-decision map ⁇ (s), k mixed (x) produces a hard decision if the SINR of a k th user of base station (s) exceeds a threshold c (s), k . Otherwise, a soft decision is performed.
- the SINR may be estimated with a time-averaged error-vector measurement (EVM). Time averaging may cause a block of symbols to share the same SINR estimate.
- EVM time-averaged error-vector measurement
- An alternative mixed-decision map ⁇ (s), k mixed (x) may act on individual symbols
- Equation ⁇ ⁇ 35 Equation ⁇ ⁇ 35 where the constellation space for the symbol of a k th user of base station (s) is partitioned into hard- and soft-decision regions with C (s), k (b) denoting the hard-decision region for a symbol b from that user's constellation.
- C (s), k (b) a hard decision for x is made.
- Alternative embodiments of the invention may employ different partitions of the constellation space. For example, edge constellation points may be given unbounded hard-decision regions.
- Both the average SINR and instantaneous approaches are applicable to any known constellation; they need not be restricted to BPSK, QPSK, or even QAM. Either of these mixed-decision approaches may be performed with the additional constraint that the receiver knows only the constellation employed for a subset of the active codes. Such situations may arise in EV-DO and HSDPA networks. In such cases, the receiver may use soft decisions for codes employing an unknown modulation. Those skilled in the art will understand that a modulation classification of these codes may be performed, which may be particularly useful in systems wherein all interfering codes share the same unknown constellation.
- the following algorithm which is illustrated in FIG. 13 , demonstrates one embodiment for performing IIC.
- FIG. 13 shows an internal feedback loop comprising operations 1308 , 1301 , 1302 , 1306 , and an external feedback loop comprising operations 1308 , 1301 , 1303 , and 1304 .
- the output of the external feedback loop ⁇ acute over (q) ⁇ R ⁇ [i] ⁇ circumflex over (b) ⁇ [i] , which is multiplicatively scaled 1305 by ⁇ [i] .
- the scaled output is combined 1306 with the internal feedback loop to yield ( q ⁇ R ⁇ [i] ⁇ circumflex over (b) ⁇ [i] )+F ⁇ [i] ⁇ circumflex over (b) ⁇ [i] , which is processed by a symbol estimator 1307 and fed to the iteration delay 1308 that begins the internal and external loops.
- embodiments of the invention are described with respect to forward-link channels, embodiments may be configured to operate in reverse-link channels.
- reverse link different users' transmissions experience different multipath channels, which requires appropriate modifications to Rake processing and signal synthesis.
- a front-end processor may incorporate one Rake for every user in every base station rather than a single Rake per base station.
- a separate multipath channel emulator may be employed for imparting multipath delays and gains to each user's signal. Accordingly, the number of constituent finger signals will equal the sum over the number of multipath fingers per user per base station, rather than the sum over the number of multipath fingers per base station.
- ASICs Application Specific Integrated Circuits
- FPGAs Field Programmable Gate Arrays
- DSPs Digital Signal Processors
- Software and/or firmware implementations of the invention may be implemented via any combination of programming languages, including Java, C, C++, MatlabTM, Verilog, VHDL, and/or processor specific machine and assembly languages.
- Computer programs i.e., software and/or firmware implementing the method of this invention may be distributed to users on a distribution medium such as a SIM card, a USB memory interface, or other computer-readable memory adapted for interfacing with a consumer wireless terminal.
- a distribution medium such as a SIM card, a USB memory interface, or other computer-readable memory adapted for interfacing with a consumer wireless terminal.
- computer programs may be distributed to users via wired or wireless network interfaces. From there, they will often be copied to a hard disk or a similar intermediate storage medium.
- the programs When the programs are to be run, they may be loaded either from their distribution medium or their intermediate storage medium into the execution memory of a wireless terminal, configuring an onboard digital computer system (e.g., a microprocessor) to act in accordance with the method of this invention. All these operations are well known to those skilled in the art of computer systems.
Abstract
Description
with the following definition
-
- (0, T) is the symbol internal;
- B is the number of modeled base stations and is indexed by the subscript (s) which ranges from (0) to (B−1); here, and in the sequel, the term “base stations” will be employed loosely to include cells or sectors;
- L(s) is the number of resolvable (or modeled) paths from base station (s) to the mobile;
- α(s), l and τ(s), l are the complex gain and delay, respectively, associated with the l-th path of base station (s);
- K(s) is the number of active users or subchannels in base station (s) that share a channel via code-division multiplexing; these users or subchannels are indexed from 0 to K(s)−1;
- u(s), k(t) is a code waveform (e.g., spreading waveform) of base station (s) used to carry the kth user's symbol for the base station (e.g., a chip waveform modulated by a user-specific Walsh code and covered with a base-station specific PN cover);
- b(s), k is a complex symbol transmitted for the kth user or subchannel of base station (s);
- and w(t) is zero-mean complex additive noise that contains both thermal noise and any interference whose structure is not explicitly modeled (e.g., inter-channel interference from unmodeled base stations and/or intra-channel interference from unmodeled paths).
The advanced signals are scaled 212-213 by corresponding path gains
prior to combining 214 to produce a combined signal of the form
where
is the Euclidean norm of the path-gain vector,
and the superscript T denotes the matrix transpose operator.
This value is also referred to as a Rake front-end soft estimate of the symbol b(s), k. Since Rake processing, combining, and despreading are linear operations, their order may be interchanged. Thus, alternative embodiments may be provided in which the order of the linear operations is changed to produce q(s), k.
1 of transmitted symbols
produced by an ith ICU is input to scaling
to produce weighted symbol estimates
The magnitude of weight γ(s), k [i] may be calculated with respect to a merit of the corresponding symbol estimate {circumflex over (b)}(s), k [i].
where SINR(s), k [i] is s a ratio of average signal power to interference-plus-noise power of a kth user in base station (s) after the ith ICU, and Ck is a non-negative real constant that can be used to ensure some feedback of a symbol estimate, even if its SINR is small. Note that, as the SINR grows large, the weight tends toward unity, meaning that the estimate is very reliable.
where Re{ } returns the real part of the argument. The statistical expectations E[ ] in the numerator and denominator can be estimated, for example, via time-series averaging. The term slice ({circumflex over (b)}(s), k [i]) represents the symbol estimate {circumflex over (b)}(s), k [i] sliced (i.e., quantized) to the nearest constellation point from which the symbol b(s), k was drawn. This approach is applicable for symbols with known constellations. For example, it is typical for a receiver to know the symbol constellation for a user of interest, but it may not know which constellations are assigned to other users.
γ(s),k [i]=0 for some subset of the users.
(i.e., interference of all users cancelled). In some embodiments, the weights of user signals transmitted from at least one weakest base station are set to zero.
. Channel emulation (including delaying the synthesized transmission 515-516 with channel gains α(s), l) is performed to produce constituent signals corresponding to each finger. A synthesized constituent signal for an lth finger of base station (s) is
When all of the finger constituents are summed, the result is
which is an estimate of the signal that would be received at the mobile if the base stations were to transmit the weighted symbols.
The subscript k on the left-hand side denotes that the constituent signal is for a user k, whereas the subscript l on the left-hand side of
The left-hand sides of
where {tilde over (γ)}(s), j [i] is a jth constituent signal (either finger or user) for base station (s) A plurality of B of these sums corresponding to different base stations are combined in combining
The synthesized receive signal is subtracted from the actual received signal in a
z (s),j [i](t)≡{tilde over (y)} (s),j [i](t)+μ[i](y(t)−{tilde over (y)} [i](t)),
where z(s), j [i](t) is an interference-cancelled jth constituent signal for base stations (s).
where J(s) ⊂{0, 1, . . . , J(s)−1} are indices of the subset of constituent signals to be employed in cancellation. Embodiments of the invention may be configured for applications in which hardware limitations restrict the number of finger signals or user signals that can be used for interference cancellation (e.g., only the strongest constituents are used).
onto code waveforms associated with base station (s) via correlative despreading. The resulting quantity for a kth user of base station (s) is denoted by
are weighted by a plurality of weighting modules 833-834, and the weighted signals are combined in
The values {tilde over (b)}(s), k [i] shown in Equation 11 and Equation 12 are generally not the same value, since the value of {tilde over (b)}(s), k [i] in Equation 11 is produced by cancellation employing finger constituents, whereas {tilde over (b)}(s), k [i] expressed by Equation 12 is produced by cancellation employing user constituents.
The difference signal for base station (s) is processed by a parallel bank of time advance modules 912-913 associated with the multipath channel for that base station, followed by maximal-ratio combining. In this embodiment, a maximal-ratio combining module is configured to perform weighting 914-915 and combining 916. A resolving module comprising multipliers 917-918 and integrators 919-920 resolves the resulting combined signals onto code waveforms of the base station's users to give the difference signal vector, or error signal vector, q(s), k−{tilde over (q)}(s), k [i], where
in which all of the users' soft weights are ordered first by base station and then by users within a base station. The same indexing may also be used to express the column vector of symbol estimates input to an ith ICU as
The weighted symbol estimates are expressed as Γ[i] b [i], and the outputs of the Rake despreading modules 901-903 are expressed by the difference equation,
The values of q [i] represent the despread signals, such as described with respect to
is employed for ordering users (first by base station, and then by users within a base station) described with respect to
Thus, the elements of R can be built at the receiver with estimates of the path gains, path delays, and knowledge of the users' code waveforms.
{tilde over (b)} [i+1] =FΓ [i] {circumflex over (b)} [i]+μ[i]( q−{tilde over (q)} [i]).
(F (s)(s))kk′ =∫u (s),k(t)u (s),k′*(t)dt. Equation 21
are jointly complex normal random variables with mean Rb and covariance Γ[i]R (i.e., q|b is distributed as CN(Rb; R)). If it is approximated that q|{tilde over (b)} [i+1] is distributed as CN(R{tilde over (b)} [i+1]; R), where {tilde over (b)} [i+1] and its dependence on μ[i] are given by
Alternatively, the value of μ[i] that gives the maximum-likelihood soft estimate for Γ[i] {tilde over (b)} [i+1] is
to produce an estimated received signal RΓ[i] {circumflex over (b)} [i]. A
Alternatively, since the operations used to produce β [i] are linear, a difference signal y(t)−{tilde over (y)}(s) [i](t) may be produced prior to despreading, such as shown by
and the norm square of this signal is calculated 1106 to produce the denominator of Equation 25.
The signal β [i] is generated by a Rake, maximal ratio combining, and
where μ[i] is fixed for every ICU and C, p and r are non-negative constants.
d (s),k [i]=1 if the symbol constellation is unknown Equation 30
for positive real-valued constants a(s), k and c(s), k. The expression Re{●} returns the real part of its argument, and Im{●} returns the imaginary part of its argument. A hard decision map is provided when Ψ(s), k(x) slices the input so that the output is an element from the complex symbol constellation employed by the kth user of base station (s),
Ψ(s),k hard(x)=slice(x). Equation 33
The slicer quantizes its argument x to the nearest constellation symbol according to some metric (e.g., Euclidean distance), A hard decision is applicable only to those symbols whose constellations are known to the receiver.
where the constellation space for the symbol of a kth user of base station (s) is partitioned into hard- and soft-decision regions with C(s), k(b) denoting the hard-decision region for a symbol b from that user's constellation. If xεC(s), k(b), then a hard decision for x is made. One embodiment for defining C(s), k(b) is to include all points within a predetermined distance of b in the constellation space,
C (s),k(b)={x:distance(x,b)<c (s),k(b)}, Equation 36
where any distance metric may be used (e.g., |x−b|p for some p>0) and the radii c(s), k(b) over the set of constellation points b are chosen such that the hard-decision regions are non-overlapping. Alternative embodiments of the invention may employ different partitions of the constellation space. For example, edge constellation points may be given unbounded hard-decision regions.
Notation:
-
- b in Equation 22
- {circumflex over (b)} [i] is in Equation 31
- q is in Equation 17
- R is in Equation 19
- F is I or as in Equation 21
- Γ[i] is in
Equation 14 with elements defined in Equation 3-Equation 5 - μ[i] is defined in Equation 23-Equation 27
- Ψ maps each argument to a complex number to implement de-biasing as in Equation 28-Equation 30 and then symbol estimation as in Equation 32-Equation 36
Initializations:
Update Equation: {circumflex over (b)} [i−1]=Ψ{μ[i]( q−RΓ [i]{circumflex over (b)} [i])+FΓ [i] {circumflex over (b)} [i]}
Output:
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US8300745B2 (en) | 2012-10-30 |
US20130343498A1 (en) | 2013-12-26 |
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US8462901B2 (en) | 2013-06-11 |
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