WO2006028654A2

WO2006028654A2 - Systems and methods for receiving data in a wireless communication network

Info

Publication number: WO2006028654A2
Application number: PCT/US2005/028733
Authority: WO
Inventors: Ismail Lakkis
Original assignee: Pulse-Link, Inc.
Priority date: 2004-09-03
Filing date: 2005-08-11
Publication date: 2006-03-16
Also published as: WO2006028654A3; US20050201473A1; US20080107199A1

Abstract

A method of communicating over a wideband communication channel is provided. In one embodiment, a communication channel (200) is divided into a plurality of non-overlapping sub-channles (202) by dividing a single serial message (B) into a plurality of parallel messages. Each of the plurality of parallel messages are then encoded onto at least some of the plurality of sub-channles (202), and then transmitted.

Description

S¹SSTEMS ANDlWEmOrøFORRECMNTNGDATAIN A WIRELESS COMMUMCAΗONNETWORK

1. Field of 1fae Invention

The invention relates generally to wireless cαrnmunicalion and more particularly to systems and methods for wireless communication over a wide bandwidth channel using aplurality of sub-channels.

2. Background

Wireless ∞mmunication systems are proliferating at the Wide Area Network (WAN), Local Area Network (LAN), and Personal Area Network (PAN) levels. These wireless communication systems use a variety of techniques to allow simultaneous access to multiple users. The most common of these techniques are Frequency Division Multiple Access (FDMA), which assigns specific frequencies to each user, Time Division Multiple Access (TDMA), which assigns particular time slots to each user, and Code Division Multiple Access (CDMA), which assigns specific codes to each user. But these wireless communication systems and various modulation techniques are afflicted by a host of problems that limit the capacity and the quality of service provided to flie users. The following paragraphs briefly describe a few of these problems tor the purpose of illustration

One problem that can exist in a wireless communication system is multipath interference. Multipath interference, or multipath, occurs because some of the energy in a transmitted wireless signal bounces off of obstacles, such as buildings or mountains, as it travels from source to destination. The obstacles in effect create reflections of the teansmitted signal and the more obstacles there are, the more reflections they generate. The reflections then travel along their own transmission pafhs to the destination (or receiver). The reflections will contain the same information as the original signal; however, because of the differing transmission path lengths, the reflected signals will be out of phase with the original signal. As aresult, they will often combine destructively with the original signal in the receiver. This is referred to as Ming. To combat fading, current systems typically try to estimate the muMpafh effects and then compensate for them in the receiver using an equalizer. In practice, however, it is very difficult to achieve effective multipafh compensatioa

A second problem that can affect the operation of wireless communication systems is interference fiom adj acent communication cells within the system IQ FDMA/TDMA systems, this type of interference is prevent through a frequency reuse plan. Under a frequency reuse plan, available ∞mmunication frequencies are allocated to communication cells within the communication system such that the same frequency will not be used in adjacent cells. Essentially, the available fiequendes are split into groups. The number of groups is termed the reuse factor. Then the communication cells are grouped into clusters, each cluster containing the same number of cells as there are frequency groups. Each frequency group is then assigned to a cell in each cluster. Thus, if a frequency reuse factor of 7 is used, for example, then aparticular communication frequency will be used only once in every seven communication cells. Thus, in any group of seven communication cells, each cell can only use 1/7⁴¹ of the available frequencies, i.e., each cell is only able to use 1/7* of the available bandwidth.

In a CDMA communication system, each cell uses the same wideband communication channel. In order to avoid interference with adjacent cells, each communicaiion cell uses a particular set of spread spectrum codes to differentiate coramunicalions within the cell fiom those originating outside of the cell Thus, CDMA systems preserve the bandwidth in the sense that they avoid reuse planning. But as will be discussed, there are other issues that limit the bandwidth in CDMA systems as well Thus, in overcoming interference, system bandwidth is often sacrificed Bandwidth is becoming a very valuable commodity as wireless communication systems continue to expand by adding more and more users. Therefore, trading offbandwidth for system performance is a costly, albeit necessary, proposition that is inherent in all wireless communication systems.

The foregoing are just two examples of the types of problems that can affect conventional wireless communication systems. The examples also illustrate that there are many aspects of wireless communication system performance that can be improved through systems and methods that, for example, reduce interference, increase bandwidth, or both.

Not only are conventional wireless communication systems effected byproblems, such as those described in the preceding paragraphs, but also different types of systems are effected in different ways and to different degrees. Wireless communication systems can be split into three types: 1) line-of-sight systems, which can include point-to-point or point- to-multipoint systems; 2) indoor non-line of sight systems; and 3) outdoor systems such as wireless WANs. Iine-of-sight systems are least affected by the problems described above, while indoor systems are more affected, due for example to signals bouncing off of building walls. Outdoor systems are by tar the most affected of the three systems. Because these types of problems are limiting factors in the design of wireless transmitters and receivers, such designs must be tailored to the specific types of system in which it will operate. In practice, each type of system implements unique communication standards tfiat address the issues unique to the particular type of system. Even if an indoor system used the same communication protocols and modulation techniques as an outdoor system, for example, the receiver designs would still be different because multipath and other problems are unique to a given type of system and must be addressed with unique solutions. This would not necessarily be the case if cost efficient and effective methodologies can be developed to combat such problems as desαibed above that buMmprogrammabi%sothatadevirecante types of systems and still maintain superior performance.

SUMMARYOFTHEINVENΠON

In order to combat the above problems, the systems and methods described herein provide a novel channel access technology that provides acost efficient and effective methodology that builds mprogrammability so that a device can be reconfigured for different types of systems and still maintain superiorperforrnance. Ih one aspect of the invention, a meihod of communicating over a wideband-∞mmunication channel divided into a plurality of sub-channels is provided The method comprises dividing a single serial message intended for one of the plurality of communication devices into a plurality of parallel messages, encoding each of the plurality of parallel messages onto at least some of the plurality of sub-channels, and transmitting the encoded plurality of parallel messages to the communication device ova the wideband communication channel. WhM symools are resttidάPto panicular range of values, the transmitters and receivers can be simplified to eliminate high power consuming cornponents such as a local oscillator, synthesizer and phase locked loops. Thus, in one aspect a transmitter comprises aphπality of pulse αmverteisarκiαffieren^ stream into a pulse sequence which can be filtered to reside in the desired fiequency ranges and phase. The use of the balanced trinary data stream allows conventional components to be replaced by less costly, smaller components lhat consume less power.

Similarly, in another aspect, a receiver comprises detection of the magnitude and phase of the symbols, which can be achieved with an envelope detector and sign detector respectively. Thus, conventional receiver components can bereplacedbyless costly, smaUαoomponents that consume less power.

Other aspects, advantages, andnovel features ofthe invention will become apparent fiomthefollowing Detailed Description ofPreferred Embodiments, when considered in conjunction with the accompanying drawings.

BRBEFDESCRIFnQN OFTEDEDRAWINGS

Preferred embodiments of the present inventions taught herein are illustrated by way of example, and not by way oftrmitation, in the figures of ftie accompanying drawings, in which

Figure 1 is a diagram illustrating an example embodiment of a wideband channel divided into aplurality of sub¬ channels in accordance with the invention;

Figure2 is a diagram illustrating the effects ofmultipatliinawireless communication system;

Figure 3 is a diagram illustrating another example embodiment of a wideband communication channel divided into aplurality of sub-charmels in accordance with the inventioii;

Figure4κadiagramfflustratingtheappHc^onofaiOll-offfectortote 1 and2;

Figure 5A is a diagram illustrating the assignment of sub-channels for a wideband communication channel in accordance with the invention;

Figure 5B is a diagram illustrating the assignment of time slots for a wideband ∞mmunication channel in accordance with the invention;

Figure 6 is a diagram illustrating an example embodiment of a wireless communication in accordance with Hie invention;

Figure 7 is a diagram illustrating the use of synchronization codes in Hie wireless communication system of figure5 in accordance with the invention;

Figure 8 is a diagram illustrating a correlator that can be used to ∞nτelatesyncbrøriization codes in flie wireless communication system of figure 5;

Figure 9 is a diagram illustrating synchronization code correlation in accordance with the invention;

Figure 10 is a diagram illustrating the cross-correlation properties of synchronization codes configured in accordance with the invention;

Figure 11 is a diagram illustrating another example embodiment of a wireless communication system in accordance with the invention;

Figure 12A is a diagram illustrating how sub-channels of a wideband communication channel according to the present invention canbe grouped in accordance with the present invention;

Figure 12B is a diagram illustrating the assignment of the groups of sub- channels of figure 12A in accordance with the invention;

Figure 13 is a diagram illustrating the group assignments of figure 12B in the time domain;

Figure 14 is a flow chart illustrating the assignment of sub-channels based on SIR measurements in the wireless communication system of figure 11 in accordance with the invention;

Figure 15 is a logical block diagram of an example embodiment of transmiiter configured in accordance with the invention;

Figure 16 is a logical block diagram of an example embodiment of a modulator configured in accordance with 1hepresentrnventionforusein1he1ransrrdtteroffigure 15;

Figure 17 is a diagram illustrating an example embodiment of a rate controller configured in accordance with the invention fbrusein the modulator offigure 16;

Figure 18 is a diagram illustrating another example embodiment of a rate controller configured in accordance with the inventionforuseinthemodulator of figure 16;

Figure 19 is a diagram illustrating ai example embodiment of a frequency encoder configured in accordance with theinventionforuse in themodulator offigure 16;

Figure 20 is a logical block diagram of an example embodiment of a TDMZFDM block configured in accordance wititheinvention&ruseinthemodulatoroffigure 16;

Figure 21 is a logical block diagram of another example embodiment of a TDMZFDM block configured in accOrdanrewiihiheinveMonforuseinthemcK±ilatoroffigure 16;

Figure 22 is a logical block diagram of an example embodiment of a frequency shifter configured in accordance with the inventionforuse in tlαemodulator offigure 16;

Figure 23 is alogical block diagram of areceiver configured in accordance with the invention;

Figure 24 is a logical block diagram of an example embodiment of a demodulator configured in accordance with the invention for use in thereceiver of figure 23;

Figure 25 is a logical block diagram of an example embodiment of an equalizer configured in accordance with the present invention for use in the demodulator offigure 24;

Figure 26 is a logical block diagram of an example embodiment of a wireless communication device configured in accordance with the invention;

Figure 27 is a flow chart illustrating an exemplary method for recovering bandwidth in a wireless communicationnetWork in accordance with the invention; figure 28 'is a diagram ffiustfatmgWexemplary wireless communication network in which the method of figure 27 canbe implemented;

Figure 29 is a logical block diagram illustrating an exemplary transmitter that can be used in the network of figure 28 to implement the method of figure 27;

Figure 30 is a logical block diagram illustrating another exemplary transmitter that canbe used in the network of figure 28 to implementthemethod of figure 27;

Figure 31 is a diagram illustrating another exemplary wireless cx)mmuracation network in which the method of figure 27 can be implemented;

Figure 32 is a diagram illustrating a wireless communication system comprising 4 access points with overlapping coverage areas;

Figure 33 A is a diagram illustrating a wideband communication channel for use in the system of figure 32 comprising a single αmnunication band in accordance with one embodiment;

Figure 33B is a diagram illustrating a wideband communication channel for use in the system of figure 32 comprising two communication bands in accordance with one embodiment;

Figure 33C is a diagram illustrating a wideband αmmunication channel for use in the system of figure 32 comprising four communication bands in accordance with one embodiment;

Figure 34 is a diagram illustrating circuitry that canbe used in a transmitter of the system of figure 32 to generate the bands illustrated in figures 33 A-33B in accordance with one embodiment;

Figure 35 is a diagram iUustrating further cirαatry to can be used in a transmitter of the generate thebands illustrated nifigur^33A-33Bh

Figure 36 is a diagram illustrating άrcuitry tocanbeusedmatransmitterofthesystemoffigL]re32togenerate the bands illustrated in figures 33 A-33B in accordance with another embodiment;

Figure 37 is a diagram illustrating an example fiame structure to can be used to implement alow data rate mode in the system of figure 32 in accordance with one embodiment;

Figure 38 is a diagram illustrating one possible implementation of aheader included in the fiame of figure 37;

Figure 39 is a diagram illustrating one possible implementation of a dataportion of the fiame of figure 37;

Figure 40 is a diagram illustrating further circuitry to canbe included in a transmitter used in the system of figure 32 in accordance with one embodiment;

Figure41 is adiagtanfflustratmganencodertoc^te embodiment;

Figure 42 is a diagram illustrating an example encoding scheme to can be used in a transmitter used in the system of figure 32;

Figure 43 is a diagram iUustrating a wideband channel corriprisingmulφle bands for use in the system of figure 32 in accordance with one embodiment; "Figure 44 is a diagram illustrating αrcuitty that can be used in a transmitter use din fee system of figure 32 in accordance with one eambodiment;

Figure45 isadiagramillustratingalookup table that canbeincludedmthecinMtiy of figure 44; and

Figure 46 is a diagram illustrating circuitry that can be used in a transmitter use din the system of figure 32 in accordance wiflti another embodiment

DETAILED DESCEIPπON OFTHE PREEEJtREDEVIBODIMEINTS 1. Introduction

In order to improve wireless communication system peribrraance and allow a single device to move fiorn one type of system to another, while still maintaining superior performance, the systems aid methods described herein provide various communication methodologies that enhance performance of transmitters and receivers with regard to various common problems that afflict such systems and that allow the transmitters and/or receivers to be reconfigured for optimal performance in a variety of systems. Accordingly, the systems and methods described herein define a channel access protocol that uses a common wideband communication channel for all communication cells. The wideband channel, however, is then divided into aplurality of sub-channels. Different sub-channels are then assigned to one or more users within each cell But the base station, or service access point, within each cell transmits one message that occupies the entire bandwidth of the wideband channel. Each user's communication device receives the entire message, but only decodes those portions of the message that reside in sub-channels assigned to the user. For a point-to-point system, for example, a single user may be assigned all sub-channels and, therefore, has the full wide band channel available to them Ih awireless WAN, on the other hand, the sub-channels maybe divided among aplurality of users.

In the descriptions of example embodiments that follow, implementation differences, or unique concerns, relating to different types of systems will be pointed outtothe extent possible. But it shouldbeundeistoodtriatihesys^ and methods described herein are applicable to any type of communication systems. In addition, terms such as communication cell, base station, service access point, etc. are used interchangeably to refer to the common aspects of networks at these different levels. To begin illustrating the advantages of the systems and methods described herein, one can start by looking at the multipath effects for a single wideband communication channel 100 of bandwidth B as showninfigure 1. Communications sent ova charniel lOOmairaditionalwMesscommuricatimsysto digital databits, or symbols, that are encoded andmodulated onto aRF canierthatis centered at frequency^ and occupies bandwidthfi Generally, the width ofthe symbols (or the symbol duration) Tis defined as i^S. Thus,ifthebandwidth5is equal to lOOMHz, then the symbol duration Tis definedby the following equation:

T= 1/B = 1/100 megahertz (MHZ) = 10 nanoseconds (ns). (1)

When a receiver receives Hie communication, demodulates it, and then decodes it, it will recreate a stream 104 of data symbols 106 as illustrated in figure 2. But the receiver will also receive multipath versions 108 ofthe same data stream Because multipath data streams 108 are delayed in time relative to file data stream 104 by delays dl, d2, d3, and d4, for example, theymayoDn±)ine destructively with data stream 104. A delay spread cζ is α&fined'as the'delay irom recφtion of data stream 104 to the reception of the last multipaih data stream 108 that interferes with the reception of data stream 104. Thus, in the example illustrated in figure 2, the delay spread d_s is equal to delay d4. The delay spread d_s will vary for different environments. An environment with a lot of obstacles will create a lot of multipaήi reflections. Thus, the delay spread d_s will be longer. Experiments have shown that for outdoor WAN type environments, the delay spread 4 can be as long as 20 microseconds. Using Hie 10ns symbol duration of equation (1), ftiis translates to 2000 symbols. Thus, with a very large bandwidth, such as 100MHz, multipath interference can cause a significant amount of interference at the symbol level for which adequate compensation is difficult to achieve. This is true even for indoor environments. For indoor LAN type systems, the delay spread d_s is significantly shorter, typically about 1 microsecond. For a 10ns symbol duration, ftiis is equivalent to 100 symbols, which is more manageable but still significant By segmenting the bandwidth B into a plurality of sub-channels 202, as illustrated in figure 2, and generating a distinct data stream for each sub-charmel,1hemultipaih effect ran be reduced to a much more manageable level. For example, if the bandwidth b of each sub-channel 202 is 500KHz, then the symbol duration is 2 microseconds. Thus, the delay spread d_s for each sub-channel is equivalent to only 10 symbols (outdoor) or half a symbol (indoor). Thus, by breaking up amessage that occupies the entire bandwidflii? into discrete messages, each occupying Hie bandwidth b of sub-channels 202, a very wideband signal tot suffers from relatively minor multipath effects is created

Before discussing further features and advantages of using a wideband communication channel segmented into a plurality of sub-channels as desαribed, certain aspects of the sub-channels will be explained in more detail Referring backto figure 3, the overall bandwidth!? is segmented into iVsub-channels center at f

requencies^ Thus,thesub- channel 202 that is immediately to the right of/e is offset from_/e by b/2, where & is the bandwidlh of each sub-channel 202. The next sub-channel 202 is offset by 3b/2, the next by 5b/2, and so on. To the left of/e, each sub-channel 202 is offsetby -b/2, -3b/2, -5b/2, etc.

Preferably, sub-channels 202 are non-overlapping as this allows each sub-channel to be processed independently in the receiver. To accomplish this, a roll-off factor is preferably applied to the signals in each sub-channel in apulse-shaping step. The effect of such apulse-shaping step is illustrated in figure 3 by the non-rectangular shape of the pulses in each sub-channel 202. Thus, the bandwidlh b of each sub-cliannelcanberepresentedbyanequationsuchasthe following: b = (l+r)/T; (2)

T= the symbol duration.

Without the roll-off factor, i.e., b = IfT, the pulse shape would be rectangular in the frequency domain, which correspondsto a(fø^Λfunctionin1he time domain The time domain signal fora^^)Λ; signal 400 is shown in figure 4 in order to illustrate theproblems associated with arectangularpulse shape andtheneed touse aroU-off factor.

As can be seen, main lobe 402 comprises almost all of signal 400. But some of the signal also resides in side lobes 404, which stretch out indefinitely in both directions from main lobe 402. Side lobes 404 make processing signal 400 muc&'riibre difficult, which increases ffiie complexity of the receiver. Applying a roll-off factor r, as in equation (2), causes signal 400 to decay faster, reducing the number of side lobes 404. Thus, increasing the roll-off factor decreases the length of signal 400, Le., signal 400 becomes shorter in time. But including the roll-off factor also decreases the available bandwidth in each sub-channel 202. Therefore, r must be selected so as to reduce the number of side lobes 404 to a suffirientnumber, e.g., 15, wMestfflmaximizingihe available bandwidJhin each sub-channel 202.

Thus, the overall bandwidth^ tor communication channel 200 is givenby the following equation:

B=N(l+r)/T; (3) or

B=MfT; (4)

^"Wbae

M=(l+r)N. (5)

For efficiency purposes related to transmitter design, it is preferable Ihatz-is chosen so thatMin equation (5) is an integer. Choosing r so thatMis an integer allows for more efficient transmitters designs using, for example, Inverse Fast Fourier Transform (ik¥ 1) techniques. SnaceM=N+N(r), andNisalwaysaiiinteger,1hismeam1hatrmustbechosen so thatiVfrJ is an integer. Generally, it is preferable forrtobe between 0.1 and 0.5. Therefore, ifNis 16, for example, then 0.5 could be selected for r so ihstN(r) is an integer. Alternatively, if a value for r is chosen in the above example so that N(r) is not an integer, B can be made slightly wider than MT to compensate. In this case, it is still preferable that r be chosen so thatiVf?;) is approximately an integer.

2. Example Embodiment of a Wireless Communicalion System

With the above in mind, figure 6 illustrates an example communication system 600 comprising a plurality of cells 602 that each use a common wideband cmimumcafion channel to communicate with communication devices 604 within each cell 602. The common communication channel is a wideband αarnmunication channel as described above. Each communication cell 602 is defined as the coverage area of a base station, or service access point, 606 within the cell One such base station 606 is shown for illustration in figure 6. For purposes of this specification and the claims that follow, the term base station will be used generically to refer to a device that provides wireless access to the wireless communication system for a plurality of communication devices, whether the system is a line of sight, indoor, or outdoor system.

Because each cell 602 uses the same rømmunication channel, signals in one cell 602 must be distinguishable from signals in adjacent cells 602. To differentiate signals from one cell 602 to another, adjacent base stations 606 use different synchronization codes according to a code reuse plan In figure 6, system 600 uses a synchronization code reuse factor of 4, although the reuse factor can vary depending on ftte application.

Preferably, the synchronization code is periodically inserted into a rorrmurήcation from a base station 606 to a communication device 604 as illustrated in figure 7. Aflerapredetemimednurnberofdatapackets7Q2,intecasetwo, the particular synchronization code 704 is inserted into the information being transmitted by each base station 606. A synchtOnization code is a sequence olidatabϊtsiαiown to botfi fee base station 606 and any communication devices 604 with which it is communicating. The synchronization code allows such a communication device 604 to synchronize its timing to that ofbase station 606, which, in turn, allows device 604 to decode 1hedatapiυperly.Thus,incell l

shaded cells 602 in figure 6), for example, synchronization code 1 (SYNCl) is inserted into data stream 706, which is generated by base station 606 in cell 1, after every two packets 702; in cell 2 SYNC2 is inserted after every two packets 702; in cell 3 SYNC3 is inserted; and in cell 4 SYNC4 is inserted. Use of flie synchronization cedes is disαissed ;in more detail below.

In figure 5A, an example wideband communication channel 500 tor use in communication system 600 is divided into 16 sub-channels 502, centered at fiequendes fo to

A base station 606 at the center of each communication cell 602 transmits a single packet occupying the whole bandwidth B of wideband channel 500. Such a packet is illustrated bypacket 504 in figure 5B. Packet 504 comprises sub-packets 506 that are encoded with a frequency offset corresponding to one of sub-channels 502. Sub-packets 506 in effect define available time slots in packet 504. Similarly, sub-channels 502 can be said to define available frequency bins in communication channel 500. Therefore, the resources available in communication cell 602 are time slots 506 and frequency bins 502, which can be assigned to different communication devices 604 within each cell 602. Thus, for example, frequency bins 502 and time slots 506 can be assigned to 4 different communication devices 604 wilhin a cell 602 as shown in figure 5. Each communication device 604 receives the entire packet 504, but only processes those frequency bins 502 and/or timeslots 506 that are assigned to it Preferably, each device 604 is assigned non-adjacent frequency bins 502, as in figure 5B. This way, if interference corrupts the information in a portion of communication channel 500, then the effects are spread across all devices 604 within a cell 602. Hopefully, by spreading out the effects of interference in this manner the effects are minimized and Hie entire infbmiation sent to each device 604 can still be recreated from the unaffected information received in other frequency bins. For example, if interference, such as fading, corrupted the information in \msj₀-f₄ then each user 14 loses one packet of data. But each user potentially receives three unaffected packets from the other bins assigned to them. Hopefully, the unaffected data in the oiher three bins provides enough Mor^ message for each user. Thus, frequency diversity can be achieved by assigning non-adjacent bins to each of multiple user's.

Ensuring Ihat the bins assigned to one user are separated by more than the coherence bandwidth ensures frequency diversity. As discussed above, the coherence bandwidth is approximately equal to IZd₉ For outdoor systems, where ds is typically 1 microsecond, IZd₃ = 1/1 microsecond = 1 Mega Hertz (MHz). Thus, the non-adjacent frequency bands assigned to a user are preferably separated by at least IMHz. It is even more preferable, however, if the coherence bandwidth plus some guard band to ensure sufficient frequency diversity separate the non-adjacent bins assigned to each user. For example, it is preferable in certain implementations to ensure that at least 5 times the coherence bandwidth, or 5MHz in the above example, separates the non-adjacent bins.

Another way to provide frequency diversity is to repeat blocks of data in frequency bins assigned to aparticular user feat are separated by more than trie coherence oandwidth. Ih other words, if 4 sub-channels 202 are assigned to a user, then data block a can be repeated in the first and third sub-channels 202 and data block b can be repeated in the second and fourth sub-channels 202, provided the sub-channels are sufficiently separated in frequency. In this case, the system can be said to be using a diversity length factor of 2. The system can similarly be configured to implement other diversity lengths, e.g., 3, 4,..., /.

It shouldbe noted that spatial diversity can also be included depending on the embodiment Spatial diversity can comprise transmit spatial diversity, receive spatial diversity, or both In transmit spatial diversity, the transmitter uses a plurality of separate transmitters and a plurality of separate antennas to transmit each message. In other words, each transrrώter transmits the same message inparalleL The messages are thenreceived from the transmitters and combinedin the receiver. Because the parallel transmissions travel different paths, if one is affected by fading, the others will likely not be affected. Thus, when they are<xra±)inedintherecdver,1hemessage should berecoverableeven othαtransrrώsionpaths experienced severe fading.

Receive spatial diversity uses a plurality of separate receivers and a plurality of separate antennas to receive a single message. If an adequate distance separates the antennas, then Ihetransrrrissionpathforthesignalsreceivedbythe antennas will be different Again, this differencein the transmission paths wiU provide imperviousness to fedrng when the signals fiomthereceivers are combined. Transmit and receive spatial diversity can also be combined within a system such as system 600 so that two antennas are used to transmit and two antennas are used to receive. Thus, each base station 606 transmitter can include two antennas, for transmit spatial diversity, and each communication device 604 receiver can include two antennas, forreceive spatial diversity. If onrytransmit spatial diversity is implemented in system 600, then it can be implemented in base stations 606 or in communication devices 604. Similarly, if only receive spatial diversity is included in system 600, then it canbe implemented inbase stations 606 or cxmimunication devices 604.

The number of communication devices 604 assigned frequency bins 502 and/or time slots 506 in each cell 602 is preferably programmable in real time. Ih other words, the resource allocation within a communication cell 602 is preferably programmable in the face of varying external conditions, Le., multipath or adjacent cell interference, and varying requirements, i.e., bandwidth requirements for various users within the celL Thus, if user 1 requires the whole bandwidth to download alarge video file, for example, then the allocation ofbins 502 canbe adjusttoprovideuser 1 with more, or even all, ofbins 502. Once user 1 no longer requires such large amounts ofbandwidth, the allocation ofbins 502 canbe readjusted among all ofusers 14.

It should also be noted that all of the bins assigned to a particular user canbe used for both the forward aid reverse link Alternatively, some bins 502 canbe assigned as the forward link and some canbe assigned for use on the reverse link, depending on the implementation.

To increase capacity, the entire bandwidth^ is preferably reused in each communication cell 602, with each cell 602 being differentiatedby aunique synchronization code (see discussionbelow). Thus, system 600 provides increased irrmunity to multipath and fading as well as mcreasedbandwidrhduetoteelirm 3. SVricnronization

Figure 8 illustrates an example embodiment of a synchronization code correlator 800. When a device 604 in cell 1 (see figure 6), tor example, receives an incoming communication fiom the cell 1 base station 606, it compares the incoming data with SYNQ in correlator 800. Essentially, the device scans the incoming data trying to correlate the data with the known synchronization code, in this case SYNQ Once correlator 800 matches the incoming data to SYNQ it generates a correlation peak 804 at ftie output Multipath versions of the data will also generate correlation peaks 806, although thesepeaks 806 are generally smallerthancorrelationpeak804. ThedeΛά∞canihenusethecorrelationpeaksto perform channel estimation, which allows the device to adjust for the multipath using an equalizer. Thus, in cell 1, if correlator 800 receives a data stream comprising SYNCl, it will generate correlation peaks 804 and 806. If, on the other hand, the data stream comprises SYNC2, for example, then no peaks will be generated and the device will essentially ignore the incoming communication

Even though a data stream that comprises SYNC2 will not create any correlation peaks, it can create noise in correlator 800 that canprevent detection of correlation peaks 804 and 806. Several steps canbe taken to prevent this from occurring. One way to minimize the noise created in correlator 800 by signals fiom adjacent cells 602, is to configure system 600 so that each base station 606 transmits at the same time. This way, 1he synchronization codes can preferably be generated in such a manner that only the synchronization codes 704 of adjacent cell data streams, e.g., streams 708, 710, and 712, as opposed to packets 702 within those streams, will interfere with detection of the correct synchronization code 704, e.g., SYNCl. The synclironization codes canthenbe further configured to eliminate orreducethemterference.

For example, the noise or interference caused by an incorrect synchronization code is a function of fhe cross correlation of that synchronization code with respect to the correct code. The better the cross correlation between the two, the lower the noise level. When the cross correlation is ideal, then the noise level will be virtually zero as illustrated in figure 9 by noise level 902. Therefore, a preferred embodiment of system 600 uses synclτronization codes that exhibit ideal cross correlation, i.e., zero. Pteferably, the ideal cross correlation of ihe synchronization codes covers aperiod /that is sufficient to allow accurate detection of multipath 906 as well as multipath correlationpeaks 904. This is important so that accurate channel estimation and equalization can take place. Outside of period /, the noise level 908 goes up, because the data in packets 702 is random and will exhibit low cross correlation with the synchronization code, e.g., SYNCl. Preferably, period / is actually slightly longer then the multipath length in order to ensure that the multipath can be detected a Synchronization code generation

Conventional systems use orthogonal codes to achieve cross correlation in correlator 800. In system 600 for example, SYNCl, SYNC2, SYNC3, and SYNC4, corresponding to cells 14 (see lightly shaded cells 602 of figure 6) respectively, will all need to be generated in such a manner that they will have ideal cross correlation with each other. Ih one embodiment, if the data streams involved comprise high and low data bits, then the value "1" canbe assigned to the high databits and "-1" to the low databits. Orthogonal data sequences are then those that producea'O" output when they are exclusively ORed (XORed) together in correlator 800. The following example illustrates this point for orthogonal sequences 1 and2: sequence 1: 11-1 1 sequence 2: U 1-1

1 1-1-1 =0 Thus, when the results ofXQRing eachbit pair are added, the result is "0".

But in system 600, for example, each code must have ideal, or zero, cross correlation with each of the other codes used in adjacent cells 602. "Therefore, in one example embodiment of a method for generating synchronization codes exhibiting the properties described above, the process begins by selecting a "perfect sequence" to be used as the basis for the codes. A perfect sequence is one that when correlated wilhitselfproducesanurriber equal to fee number of bite in the sequence. For example:

Perfect sequence 1 : 11-11 11-11 1111=4 But each time a perfect sequence is cyclically shifted by one bit, the new sequence is orthogonal wife, ftie original sequence. Thus, for example, ifperfect sequence 1 is cyclically shifted by one bit and flien correlated with the original, the correlationproduces a "0" as in the following example;

Perfect sequence 1: 11-1 1 11 1-1 11 -1-1 =0 If the perfect sequence 1 is again cyclically shifted by one bit, and again correlated with the original, then it will produce a "0". Si general, you can cyclically shift aperfect sequence by any number ofbits up to its length and correlate the shifted sequence with the original to obtaina "0".

Once aperfect sequence of the correct length is selected, the first synchronization code is preferably generated in oneenix)άtaeotbyrepeatingthesequence4times. Thus, ifperfect sequence 1 is being used, then a first synchronization code_y wouldbethe following y=ll-l l 11-11 11-11 11-11.

Qr in generic form: y=x(0)x(lM2)x(3)x(0)x(l)x(2)x(3)x(0)x(l)x(2K3)x(0)x(l)x(2)x(3). For a sequence oflengihL: y=x(0)x(l)...;x(L)x(0)x(i)...x(Lpi(0)κ(J)...Jc(L)x(0)xφ...x(L) Repeating theperfect sequence allows correlator 800 abetter opportunity to detect fee synchronization code and allows generation of other uncorrelated frequencies as well Repeating has the effect of sampling in the frequency domain. This effect is illustrated by the graphs in figure 10. Thus, in TRACE 1, which corresponds to synchronization code;;, a sample 1002 is generated every fourth sample bin 1000. Each sample bin is separated by 1/(4LxI), where Tis the symbol duration. Thus, in the above example, where!, = 4, each sample binis separated by 1/(16x1) in the frequency domain. TRACES 2-4 illustrate the next three synchronization codes. As can be seen, the samples for each subsequent synchronization code are shifted by one sample bin relative to the samples for the previous sequence. Therefore, none of the sequences interfere with each other.

To generate the subsequent sequences, correspondingto TRACES 2-4, sequence;^; must be shifted in frequency. This can be accomplished using 1he following equation:

I(m) =y(m)*eφ(j*2 *iήr*ni/(n *L)), (5) for r = 1 to L (# of sequences) andm= 0 to 4*L-l (time); and where: £(m) = each subsequent sequence; y(m) = the first sequence; and n=the number oftimesthe sequence is repeated

sequence is repeated n multiplied by the length of Ihe underlying perfect sequence L, in the time domain results in a shift in the frequency domain. Iiquafion(5) results in the desired shift as fflustratEd m figure 9 for each of synchrcdzatiαi codes 2-4, relative to synchronization code 1. The final step in generating each synchronization code is to append Hie copies of Hie lastMsamples, where Mis the length of file multipath, to fiie front of each code. This is done to make file convolution wifii Ihe muMpath cyclic and to allow easier detection of the multipalh.

It should be noted fliat synchronization codes can be generated from more ton one perfect sequence using the same methodology. For example, a perfect sequence can be generated and repeated four times and fiien a second perfect sequence can be generated and repeated four times to get an factor equal to eight The resulting sequence can then be sniffed as described above to create the synchronization codes, b. Signal Measurements Using Synchronization Codes

Therefore, when a communication device is at the edge of a cell, it will receive signals from multiple base stations and, therefore, will be decoding several synchronization codes at the same time. This can be illustrated with the help of figure 11, which illustrates anoftier example embc>dment of a wireless αmmumcati

communication cells 1102, 1104, and 1106 as well as communication device 1108, which is in communication with base station 1110 of cell 1102 but also receiving communication from base stations 1112 and 1114 of cells 1104 and 1106, respectively. If communications from base station 1110 comprise synchronization code SYNTCl and communications from base station 1112 aid 1114 comprise SYNC2 and SYNC3 respectively, then device 1108 will effectively receive the sum of these three synchronization codes. This is because, as explained above, base stations 1110, 1112, and 1114 are configured to transmit at the same time. Also, tiie synchronization codes arrive at device 1108 at almost file same time because they are generated in accordance with the description above.

Again as described above, Hie synchronization codes SYNCl, SYNC2, and SYNC3 exhibit ideal cross coπelation. Therefore, when device 1108 correlates the sumx of codes SYNCl, SYNC2, and SYNC3, the latter two will not interfere with proper detection of SYNCl by device 1108. Importantly, the sum* can also be used to determine important signal characteristics, because the sum x is eqi^toihesumofthesynchTOriization∞des with file following equation: x =SmCl + SYNC2 + SYNC3. (6) Therefore, when SYNCl is removed, the sum of SYNC2 and SYNC3 is left, as shown in the following: x-SYNCl =SYNC2+SYNC3. (7)

The energy computed from the sum (SYNC2 + SYNC3) is equal to the noise or interference seen by device 1108. Since thepurposeofconelatingthe synchronization codein device llOόistoextracttheenergyinSYNC 1, device 1108 also has the energy in the signal from base station 1110, i.e.,1heenergyiBpresentedby SYNCl. Therefore, device 1106 can use the energy of SYNCl and of (S YNC2 + S YNC3) to perform a signal-to-interference measurement for the communication channel over which it is communicating with base station 1110. The result of the measurement is preferably a agnal-to-interference ratio (SIR). The SIR measurement can then be communicated back to base station 1110 for purposes that will be discussed below. The ideal cross correlation of the synchronization codes, also allows device 1108 to perform extremely accurate determinations of the Channel Impulse Response (CIR), or channel estimation, from the correlation produced by correlator 800. This allows for highly accurate equalization using low cost, low complexity equalizers, thus oveicoming a significant drawback of conventional systems. 4. Sub-Channel Assignments

As mentioned, the SIR as deterrninedby device 1108 canbe ccαnmunicatedbackto base station lllOforuse in the assignment of channels 502. Ih one embodiment, due to the fact that each sub-channel 502 is processed independently, the SIR for each sub-channel 502 can be measured and communicated back to base station 1110. In such an embodiment, therefore, sub-channels 502 can be divided into groups and a SIR measurement for each group can be sent to base station 1110. This is illustrated in figure HA, which shows a wideband communication channel 1200 segmented into sub-channels ^o tofa Sub-channels/? toβs are then grouped into 8 groups Gl to G8. Thus, in one embodiment, device 1108 andbase station 1110 communicate ova achaπnel such as channel 1200.

Sub-channels in the same group are preferably separated by as many sub-channels as possible to ensure diversity. In figure 12A for example, sub-channels within the same group are 7 sub-channels apart, e.g., group Gl comprises^ andjs. Device 1102 reports a SIRmeasurement for each of the groups Gl to G8. These SIRmeasurements are preferably compared with a threshold value to determine which sub-channels groups are useable by device 1108. This comparison can occur in device 1108 or base station 1110. If it occurs in devi∞ 1108, thm device 1108 CHisirrrply report tobase station 1110 which sub-channels groups are useable by device 1108.

SIR reporting will be simultaneously occurring for a plurality of devices within cell 1102. Thus, figure 12B illustrates the situation where two communication devices corresponding to User 1 and User 2 report SIR levels above the threshold for groups Gl, G3, G5, and G7. Base station 1110 preferably then assigns sub-charrnel groups to User 1 and User 2 based on the SIRreporting as illustrated inFigure 12B. When assigningthe "good" sub-channel groups to User 1 and User 2, base station 1110 also preferably assigns them based on the principles of frequency dversity. In figure 12B, therefore, User 1 andUser 2 are alternately assigned every other "good" sub-chaπneL

The assignment of sub-channels in the frequency domain is equivalent to the assignment of time slots in the time domain. Therefore, as illustrated in figure 13, two users, User 1 and User 2, receive packet 1302 transmitted over communication channel 1200. Figure 13 also fflustratedte sub-channel assignment of figure 12B. While figures 12 and 13 illustrate sub-channel/time slot assignment based on SIR for two users, the principles illustrated can be extended for any number of users. Thus, a packet within cell 1102 can be received by 3 or more users. Although, as the number of available subchannels is reduced due to high SIR, so is the available bandwidth. In other words, as available channels are reduced, the number of users that can gain access to ccmmuricaticn channel 1200 is also reduced

Poor SIR can be caused for a variety of reasons, but frequently it results fiom a device at the edge of a cell receiving ∞mmunicatioii signals fiom adjacent cells. Because each cell is using the same bandwidth^, the adjacent cell signals will eventually raise the noise level and degrade SIR for certain sub-channels. In certain ernbodimeπts, therefore, sub-channel assignment can be coordinated between cells, such as cells 1102, 1104, and 1106 in figure 11, in order to prevent interference from adjacent cells.

Thus, if rømmunication device 1108 is near the edge of cell 1102, and device 1118is near the edge of cell 1106, then the two can interfere with each other. As a result, the SIR measimnentstø device 1108 and 1118 report back to base stations 1110 and 1114, respectively, will indicate that the interlα^nce level is tco high. Base station 1110 can then be configured to assign only the odd groups, i.e., Gl, G3, G5, etc., to device 1108, while base station 1114 can be configured to assign the even groups to device 1118. The two devices 1108 and 1118 will then not interfere with each other due to the coordinated assignment of sub-channel groups.

Assigning the sub-channels in this manner reduces the overall bandwidth available to devices 1108 and 1118, respectively. Ii this case the bandwidth is reduced by a factor ofiwo. But it shoiM be rememberaitø closer to each base station 1110 and 1114, respectively, will still be able to use all channels if needed Thus, it is only devices, such as device 1108, that are near the edge of a cell that will have the available bandwidth reduced (Contrast this with a CDMA system, for example, in which the bandwidth for all users is reduced, due to the spreading techniques used in such systems, by approximately a factor of 10 at all times. It ran be seen, therefore, that the S3«tems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels not only improves the quality of service, but can also increase the available bandwidth agrrificantly.

When there are three devices 1108, 1118, and 1116 near the edge oftfieir respective adjacent cells 1102, 1104, and 1106, the sub-channels canbe dividedby three. Thus, device 1108, for example, canbe assigned groups Gl, G4, etc, device 1118 canbe assigned groups G2, G5, etc, and device 1116 can be assigned groups G3, G6, etc. Ih this case the available bandwidth for these devices, i.e, devices near the edges of cells 1102, 1104, and 1106, is reduced by afactor of 3, but this is still better thanaCDMA system, for example.

The manner in which such a coordinated assignment of sub-charmelscanworkisillustratedbylhe flow chart in figure 14. First in step 1402, a communication device, such as device 1108, reports the SIR for all sub-channel groups Gl to G8. The SIRs reported are then compared, in step 1404, to a threshold to determine if the SIR is sufficiently low for each group. Alternatively, device 1108 canmake Hie deteimination and simply report which groups are above or below Hie SlR threshold If the SIR levels are good for each group, then base station 1110 can make each group available to device 1108, instep 1406. Periodically, device 1108 preferably measures the SIR level and updates base station 1110 in case 1he SIR as deteriorated. For example, device 1108 may move fom near the center of cell 1102 toward 1he edge, where interference fiom an adj acent cell may affect the SIR for device 1108.

If Hie comparison in step 1404 reveals that the SIR levels are not good, then base station 1110 can be preprogrammed to assign either the odd groups or the even groups only to device 1108, which it will do in step 1408. Device 1108ihenreportetheSIRmeasurementsforflieoddOTevengroupsitisassigαedinstφ 1410, and they are again comparedto aSIRIhresholdinstep 1412.

Sis assumed that the poor SIR level is due to the feet that device 1108 is operating atthe edge ofcell 1102 and is therefore being interfered with by a device such as device 1118. But device 1108 will be interfering wilh device 1118 at the same time. Therefore, the assignment of odd or even groups in step 1408 preferably corresponds with the assignment of the opposite groups to device 1118, by base station 1114. Accordingly, when device 1108 reports the SIR measurements for whichever groups, odd or even, are assigned to it, die comparison in step 1410 should reveal that the SIR levels are now below the threshold level. Thus, base station 1110 makes the assigned groups available to device 1108 in step 1414. Again, device 1108 preferably periodically updates the SIRmeasurements by returning to step 1402.

It is possible for Hie comparison of step 1410 to reveal that the SIR levels are still above the threshold, which should indicate thatathird device, e.g., device 1116 is stfflinterferingwifli device 1108. In this case, base station 1110 can be preprogrammed to assign every third group to device 1108 in step 1416. This should correspond with the corresponding assignments of non-interfering channels to devices 1118 and 1116 by base stations 1114 and 1112, respectively. Thus, device 1108 should be able to operate on the sub-channel groups assigned, Le., Gl, G4, etc., without

Optionally, a third comparison step (not shown) canbe implemented after step 1416, to ensure that the groups assigned to device 1408 posses an adequate SIR level for proper operation. Moreover, if there are more adjacent cells, i.e., if it is possible for devices in a 4^th or even a 5^fc adjacent cell to interfere with device 1108, then the process of figure 13 would continue and the sub-channel groups would be divided even further to ensure adequate SIR levels on the sub-channels assigned to device 1108.

Even though theprocess of figure 14 reduces the bandwidth available to devices atthe edge of cells 1102, 1104, and 1106, the SIRmeasurements canbe used in such a manner as to increase the data rate and therefore restore or even increase bandwidth. To accomplish this, the transmitters andieceiversusedinbase stations 1102, 1104, and 1106, andin devices in communication therewith, e.g., devices 1108, 1114, and 1116 respectively, must be capable of dynamically changing the symbol mapping schemes used for some or all of the sub-channel. For example, in some embodiments, the symbol mapping scheme canbe dynamically changed among BPSK, QPSK, 8PSK, 16QAM, 32QAM, etc. As the symbol mapping scheme moves higher, i.e, toward 32QAM, the SIR level required for proper operation moves higher, i.e., less and less interference can be withstood Therefore, once the SIR levels are determined for each group, the base station, e.g., base station 1110, can then determine what symbol mapping schaπe can be supported for each si±Kih-ttmel group and can change the modulation scheme accoidingly. Device 1108 must also change the symbol mapping scheme to correspond to that of the base stations. The change can be effected tor all groups urHibnnty, or it can be effected for individual groups. Moreover, the symbol mapping scheme can be changed on just the forward link, just the reverse link, or both, depending onthe embodiment

Thus, by maintaining the capability to dynamically assign sub-channels and to dynamically change the symbol mapping scheme used for assigned sub-channels, the systems and methods described herein provide the ability to maintain higher available bandwidths with higher performance levels than conventional systems. To fully realize Hie benefits described, however, the systems and methods described, thus far must be capable of implementation in a cost effect and convenient manner. Moreover, the irφlementation must include reconfigurability so that a single device can move between different types of communication systems and still maintain optimum performance in accordance with the systems and methods described herein The following descriptions detail example high level embodiments of hardware implementations configured to operate in accordance with the systems and methods desαibed herein in such a manner as to provide the capabilityjust described above. 5. Sample Transmitter Embodiments

Figure 15 is logical block diagram illustrating an example embodiment of a transmitter 1500 configured for wireless ∞rnmunication in accordance with the systems and methods described above. The transmitter could, for example be within a base station, e.g., base station 606, or within a communication device, such as device 604. Transmitter 1500 is provided to illustrate logical components that can be included in a transmitter configured in accordance with the systems and methods described herein It is not intended to limit the systems and methods for wireless cornmunication over a wide bandwidth channel using a plurality of sub-channels to any particular transmitter configuration or any particular wireless cornmunication system.

With this inmind, it canbe seen that transmitter 1500 rømprises a serial-to-parallel converter 1504 configured to receive a serial data stream 1502 comprising a data rate R Serial-to-parallel converter 1504 converts data stream 1502 intoNparalleldatastreams 1504, whereNisthenumberof sub-channels 202. Sshouldbenotedthatwhile the discussion that follows assumes that a single serial data stream is used, more than one serial data stream can also be used if required or desired. In any case, the datarate of each parallel data stream 1504 is theniWV Eachdatastreaml504istliensenttoa scrambler, encoder, and interleaver block 1506. Scrambling, encoding, and interleaving are common techniques implemented in many wireless communication transmitters and help to provide robust, secure cornmunication. Examples ofthese techniques will bebriefly explained for illustrativepurposes.

Scrambling breaks up the data to be transmitted in an effort to sm∞th out the sr^^ density of 1he transmitted data For example, if the data comprises a long string of Ts, there will be a spike in the spectral density. This spike can cause greater irώerference within the wireless αmmunicatimsystem3ybrealαngιφ smoothed out to avoid any suchpeaks. Often, scrambling is achievedby XORing the data with arandom sequence. Encoding, or coding, the parallel bit streams 1504 can, for example, provide Forward Error Correction (EEQ. The purpose of FEC is to improve the capacity of a communication channel by adding some carefully designed redundant information to the data being transmitted through the channel. The process of adding this redundant information is known as channel coding. ConvoMonal coding and block coding are the two major forms of channel coding. ConvoMonal codes operate on serial data, one or a few bits at a time. Block codes operate on relatively large (typically, up to acouple ofhundred bytes) message blocks. There are a variety of useful ConvoMonal andblock codes, and a variety of algorithms for decoding ftie received coded information sequences to recover the original data For example, convoMonal encoding or turbo coding with Viterbi decoding is a FEC technique that is particularly suited to a channel in which the transmitted signal is corrupted mainly by additive white gaussian noise (AWGN) or even acharmel that simply experiences fading.

ConvoMonal codes are usually described using two parameters: the code rate and the constraint length. The code rate, k/n, is expressed as a Mo of the number of bits into the convoMonal encoder (k) to the number of channel symbols (n) output by the convoMonal encoder in a given encoder cycle. A common code rate is 1/2, whichmeansthat 2 symbols are produced for every 1-bit input into the coder. The constraint length parameter, K, denotes the 'length" of the convoMonal encoder, i.e. how many M>it stages are available to feed the combinatorial logic that produces the output symbols. Closely related to K is the parameter m, which indicates how many encoder cycles an input bit is retained and used for encoding after it first appears at the input to the convoMonal encoder. The m parameter can be thought of as the memory length of the encoder.

Interleaving is used to reduce the effects of fading Interleaving mixes up the order of the data so that if a fade interferes with a portion of the transmitted signal, the overall message will not be effected This is because once the message is de-interleaved and decoded in the receiver, the data lost will comprise non-contiguous portions of the overall message. In other words, the fade will interfere with a contiguous portion of the interleaved message, but when the message is de-interleaved, the interfered with portion is spread Ihroughout the overall message. Using techniques such as FIC!, the missmg Momiatictt can tø

After blocks 1506, each parallel data stream 1504 is sent to symbol mappers 1508. Symbol mappers 1508 apply the requisite symbol mapping e.g., BPSK, QPSK, etc., to each parallel data stream 1504. Symbol mappers 1508 are preferably programmable so that the modulation applied to parallel data streams can be changed, for example, in response to the SIR reported for each sub-channel 202. It is also preferable, that each svmbolrrjarφer 1508 be sφarately programmable so that the optimum symbol mapping scheme for each sub-channel can be selected and applied to each parallel datastream 1504.

After symbol mappers 1508, parallel data streams 1504 are sent to modulators 1510. Important aspects and features of example embodiments ofmodulators 1510 are describedbelow. After modulators 1510, parallel data streams 1504 are sent to summer 1512, which is configured to sum the parallel data streams and thereby generate a single serial data stream 1518 comprising each of the individually processed parallel data streams 1504. Serial data stream 1518 is then sent to radio module 1512, where it is mόcfulated with an RF carrier, amplified, and transmitted via antenna 1516 according to known techniques.

The transmitted signal occupies the entire bandwidth i? of communication channel 100 and comprises each of the discrete parallel data streams 1504 encoded onto their respective sub-channels 102 within bandwidth B. Encoding parallel data streams 1504 onto the appropriate sub-channels 102 requires iriateachparaUel data stream 1504 be shified in βequencyby an appropriate offset This is achieved inmodulator 1510. Figure 16 is a logical block diagram of an example embodiment of a modulator 1600 in accordance with the systems and methods described herein. Importantly, modulator 1600 takes parallel data streams 1602 performs Time Division Modulation (TDM) or Frequency Division Modulation (FDM) on each data stream 1602, filters them using filters 1612, aid then shifts each data stream in frequency using frequency shifter 1614 so tet fliey occupy the appropriate sub-channel. Filters 1612 apply the required pulse shaping, Le, Hiey apply the roll-off factor described in section 1. The frequency shifts paraUel data streams 1602 are Hien summed and transmitted. Modulator 1600 can also include rate controller 1604, frequency encoder 1606, and interpolators 1610. All of the components shown in figure 16 are described in more detail in the following paragraphs andin conjunction withfigures 17-23.

Figure 17 illustrates one example embodiment of a rate controller 1700 in accordance with tie systems and methods desαibed herein. Rate control 1700 is used to control the data rate of each parallel data stream 1602. In rate controller 1700, the data rate is halved by repeating data streams d(0) to d(7), tor example, producing streams a(0) to a(l 5) in which a(0) is the same as a(8), a(l) is the same as a(9), etc. Figure 17 also illustrates that the effect of repeating the data streams in this manner is to take the data streams tet are encoded onto the first 8 sub-channels 1702, and duplicate them on Ihe next 8 sub-channels 1702. As can be seen, 7 sub-cbHmelssφaratesub-cbamiels l702∞nprising the same, or duplicate, data streams. Thus, if fading effects one sub-channel 1702, for example, the other sub-channels 1702 carrying the same data will likely not be effected, i.e., ύiere is frequency diversity between the duplicate data streams. So by sacrificing data rate, in this case half the data rate, more robust transmission is achieved. Moreover, ftre robustness provided by duplicating tie data streams d(0) to d(7) can be further enhanced by applying scrambling to the duplicated data streams via scramblers 1708.

It should be noted that the data rate can be reduced by more than half, e.g., by four or more. Alternatively, the data rate can also be reduced by an amount other ten half For example if formation from π data stream is encoded onto m sub-channels, where m >n Thus, to decrease the rate by 2/3, information from one data stream can be encoded on a first sub-channel, information from a second data stream can be erκxx3ed on a second data channel, and the sum or difference of the two data streams can be encoded on a third channel. In which case, proper scaling will need to be applied to the power in the third channel. Otherwise, for example, the power in Hie ted channel can be twice the power in the first two.

Preferably, rate controller 1700 is programmable so that the data rate can be changed responsive to certain operational factors. For example, if the SIR reported for sub-channels 1702 is low, then rate controller 1700 can be programmed to provide more robust transmission via repetition to ensure that no data is lost due to interference. Additionally, different types of wireless communication system, e.g., indoor, outdoor, line-of-sight, may require varying degrees of robustness. Thus, rate controller 1700 can be adjusted to provide the minimum required robustness for the particular type of communication system. This type of programmability not only ensures robust corrmunication, it can also be used to allow a single device to move between communication systems andmaintain superior performance.

Figure 18 illustrates an alternative example ernrxxJmentofarateconlioUer 1800 in accordance w& and methods described. Ih rate controller 1800 the data rate is increased instead of decreased This is accomplished using serial-to- parallel converters 1802 to convert each data streams d(0) to d(15), for example, into two data streams. Delay circuits 1804 then delay one of the two data streams generatedby each serial-to-parallel converter 1802 by 1/2 a symbol. Thus, data streams d(0) to d(15) are transformed into data streams a(0) to a(31). The data streams generated by a particular serial-to-parallel converter 1802 and associate delay circuit 1804 must then be summed and encoded onto the appropriate sub-channel. For example, data streams a(0) and a(l) must be summed and encoded onto the first sub¬ channel. Preferably, the data streams are summed subsequent to each data stream beingpulsed shaped by afilter 1612.

Thus, rate ∞ntroUσ l6O4 is preferably pro

1800, or decreased, as in rate controller 1700, as required by a particular type of wireless communication system, or as required by the communication channel conditions or sub-channel conditions. In the event that the data rate is increased, filters 1612 are also preferably programmable so that they can be configured to apply pulse shaping to data streams a(0) to a(31), for example, and then sum the appropriate streams to generate the appropriate nurrte of para^^ sendto fiequency shifter 1614.

The advantage of increasing the data rate in Ihe manner illustrated in figure 18 is Ihat higher symbol mapping rates can essentially be achieved, without changing the symbol mapping used in symbol mappers 1508. Once the data streams are summed, the summed streams are shifted in fiequency so triatihey reside in the appropriate sub-channeL But because the number of bits per each symbol has been doubled, the symbol mapping rate has been doubled. Thus, for example, a 4QAM symbol mapping can be converted to a 16QAM symbol mapping, even if the SIR is too high for 16QAM symbolrnappmg to oiher^^ rate in the manner illustrated in figure 18 can increase the symbol mapping even when channel conditions would otherwise not allow it, which in turn can allow a communication device to maintain adequate or even superior performance regardless of the type of communication system.

The draw back to increasing the data rate as illustrated in figure 18 is that interference is increased, as is receiver complexity. The former is due to the increased amount of data The latter is due to the feet that each symbol cannot be processed independentiy because of the 1/2 symbol overlap. Thus, these concerns must be balanced against the increase symbolmapping ability when implementing arate controller such as rate controller 1800.

Figure 19 illustrates one example embodiment of a fiequency enαxler 1900 m accordance wilhihesvstemsard methods described herein. Similar to rate encoding, fiequency encoding is preferably used to provide increased communication robustness. In frequency encoder 1900 the sum or difference of multiple data streams are encoded onto each sub-channel This is accomplished using adders 1902 to sum data streams d(0) to d(I) with data streams d(8) to d(15), respectively, while adders 1904 subtract data streams d(0) to d(7) from data streams d(8) to d(15), respectively, as shown. Thus, data streams a(0) to a(15) generated by adders 1902 and 1904 corrpise information related to mcrø than one data streams d(0) to d(15). For example, a(0) comprises the sum of d(0) and d(8), Le., d(0) + d(8), while a(8) comprises d(8) - d(0). Therefore, if either a(0) or a(8) is not received due to fading, for example, then both of data streams d(0) and d(8) can stiUbe retrieved from data stream a(8).

Essentially, 1he relationship between data stream d(0) to d(15) and a(0) to a(15) is a matrix relationship. Thus, if the receiver kmwsflierørred matrix to apply, it can recover the swosai[uaSasacesoid(0)to d(15) &am.a(0) to a(15). Preferably, frequency encoder 1900 is programmable, so that it can be enabled and disabled in order to provided robustness when required Preferable, adders 1902 and 1904 are pOgrammable also so that different matrices can be appkdtod(0)tod(15).

After frequency encoding, ifit is included, data streams 1602 are sent to TDM/FDM blocks 1608. TDMFDM blocks 1608 perform TDM or FDM on the data streams as required by the particular embodiment Figure 20 illustrates anexanpleemlx)dimer]tofaTDM/FDM^

2000 is provided to illustrate the logical components that can be included in a TDM/FDM block configured to perform TDM on a data stream. Depending on the actual implementation, some of the logical components may or may not be included TDMZFDM block 2000 comprises a sub-block repeater 2002, a sub- block scrambler 2004, a sub-block terminator 2006, asub-blockrepeater 2008, andasync inserter 2010.

Sub-block repeater 2002 is configured to receive a sub-block of data, such as block 2012 comprising bits α(0) to α(3) for example. Sub-block repeater is then configured to repeat block 2012 to provide repetition, which in turn leads to more robust communicatiαi Thus, sub-block repeater 2002 generates block 2014, which comprises 2 blocks 2012. Sub-block scrambler 2004 is then configured to receive block 2014 and to scramble it, thus generating block 2016. One method of scrambling can be to invert half ofbfock 2014 as illustrated inblock 2016. But other scrambling methods can also be implemented depending on the embodiment

Sub-block terminator 2006 takes block 2016 generated by sub-block scrambler 2004 and adds a lerrnination block 2034 to the front of block 2016 to form block 2018. Termination block 2034 ensures that each block can be prcH-essedindependentlyinthereceiver. Wi1houtterminationblock2034, scmebl∞ksmaybedelayedduetomultipath, for example, and they would therefore overlap part of thenext block of data Butbyincludingterminationblock2034₅ the delayedblockcanbe prevented fiom overlapping any of Ihe actual datainthenext block.

Termination block 2034 can be a cyclic prefix termination 2036. A cyclic prefix termination 2036 simply repeats the last few symbols of block 2018. Thus, for example, if cyclic prefix tarnination 2036 is Ifaree symbols long then it would simply repeat the last three symbols of block 2018. Alternatively, termination block 2034 can comprise a sequence of symbols that are known to both the transmitter and receiver. The selection of what type ofblock termination 2034 to use can impact what type ol equalizer is used in the receiver. Therefore, receiver complexity and choice of

After sub-bl∞kimriinator 2006, TDMZFDM block 2000 can include a sub-block repeater 2008 configured to perform asecond block repetition step in whichblock2018 is repeatedtoformblcck2020.Mcerώi embodiments, sub- block repeater can be configured to perform a second block scrambling step as well After sub-block repeater 2008, if included, TDM/FDM block 2000 comprises a sync inserter 210 configured to periodically insert an appropriate synchronization code 2032 after a predetermined number of blocks 2020 and/or to insert known symbols into each block The purpose of synchronization code 2032 is discussed in section 3.

Figure 21, on the other hand, illustrates an example embodiment of a TDM/FDM block 2100 configured for FDM, which comprises sub-block repeater 2102, sub-block scrambler 2104, block coder 2106, sub-block transformer 2108, sub-block terminator 2110, and sync inserter 2112. As with TDM/FDM block 2000, sub-block repeater 2102 repeats block 2114 and generates block 2116. Sub-block scrambler then scrambles block 2116, generating block 2118. Sub-block coder 2106 takes block 2118 and codes it, generating block 2120. Coding block correlates the data symbols together and generates symbols b. This requires joint demodulation in the receiver, which is more robust but also more complex. Sub-block transformer 2108 then performs a transfomiation on block 2120, generating block 2122. Preferably, the transformation is an IFFT of block 2120, which allows for more efficient equalizers to be used in the receiver. Next, sub-block terminator 2110 terminates block 2122, generating block 2124 and sync inserter 2112 periodically inserts a synchronization code 2126 after a certain number ofblocks 2124 and/or insert known symbols into each block Preferably, sub-block terminator 2110 only uses cyclic prefix termination as described above. Again this allows for more efficient receiver designs.

TDM/FDM block 2100 is provided to illustrate the logical components that can be included in a TDM/FDM block configured to perform FDM on a data stream Depending on the actual implementation, some of the logical components may ormaynot be included. Moreover, TDM/FDMblock2000 and2100 arepreferablyraograrnmableso that the appropriate logical components can be included as required by a particular implementation. This allows a device that incorporates one ofblocks 2000 or 2100 to move between different systems with difϊerentrequi^^ preferable that TDMZFDM block 1608 in figure 16 be programmable so that it can be programmed to perform TDM, such as described in conjunction with block 2000, or FDM, such as described in conjunction with block 2100, as requiredby aparticular communication system.

After TDM/FDMblocks 1608,infigure 15, theparallela^sitrearnsarepreferablypassed to interpolators 1610. After Interpolators 1610, the parallel data streams are passed to filters 1612, which apply the pulse shaping desαibed in conjunction with the roll-off factor of equation (2) in section 1. Then the parallel data streams are sent to frequency shifter 1614, which is configured to shift each parallel data stream by tiie frequency offset associated with the sub-channel to which theparticular parallel data stream is associated Figure 22 iltostrates an example embodiment of a f^ 2200 in accordance with the systems and methods described herein. As can be seen, frequency shifter 2200 comprises multipliers 2202 configured to multiply each parallel data stream by the appropriate exponential to achieve the required frequency shift. Each exponential is of the form: expQlif_cnTfrM), where c is the corresponding sub-channel, e.g., c = 0 to

N-I, and n is time. Preferably, frequency shifter 1614 in figure 16 is programmable so that various charmel/sub-charmel configurations can be accommodated for various different systerns. Alternatively, and filtering can be done after the IEFT block. This type of implementation can be more efficient depending on the irrplemeπtation.

After the parallel data streams are shifted, they are summed, e.g., in summer 1512 of figure 15. The summed data stream is then transmitted using the entire bandwidth B of the communication channel being used. But the transmitted data stream also comprises each of the parallel data streams shifted in frequency such that they occupy the appropriate sub-channel. Thus, each sub-channel may be assigned to one user, or each sub-channel may carry a data stream intended for different users. The assignment of sub-channels is described in section 3b. Regardless ofhow the sub¬ channels are assigned, however, each user will receive the entire bandwidth, comprising all the sub-channels, but will only decode those sub-channels assigned to the user. 6. Sample Receiver Embodiments

Figure 23 illustrates an example embodiment of a receiver 2300 that can be configured in accordance with the present invention. Receiver 2300 comprises an antenna 2302 configured to receive a message transmitted by a transmitter, such as transmitter 1500. Thus, antenna 2302 is configured to receive a wide band message comprising the entire bandwidth B of a wide band channel that is divided into sub- channels of bandwidth b. As described above, the wide bandmessage comprises aplurality of messages each encoded onto each of a rørresrrøi^ig sub-channel. Alloffhe sub-channels may or may not be assigned to a device that includes receiver 2300. Therefore, receiver 2300 may or may not be required to decode all of the sub-channels. After the message is received by antenna 2300, it is sent to radio receiver 2304, which is configured to remove the carrier associated with the wide band communication channel and extract a baseband signal comprising the data stream transmitted by the transmitter. The baseband signal is then sent to correlator 2306 and demodulator 2308. Correlator 2306 is configured to correlated with a synchronization code inserted in the data stream as described in section 3. It is also preferably ∞iifigitfed to perform SIRaM described in section 3(b). Demodulator 2308 is configured to extract the parallel data streams from each sub-channel assignedtofhe device comprising receiver 2300 andto generate a single data stream therefrom.

Figure 24 illustrates an example embodiment of a demodulator 2400 in accordance with the systems and methods described herein. Demodulator 2402 comprises a frequency shifter 2402, which is configured to apply a frequency offset to the baseband data stream so that parallel data streams comprising the baseband data stream can be independently processed in receiver 2400. Thus, the output of frequency shifter 2402 is a plurality of parallel data streams, which are then preferably filtered by filters 2404. Filters 2404 apply a filter to each parallel data stream that corresponds to the pulse shape applied in the tiHisrnitter, e.g., transmitter 1500. Alternatively, an IEFT block can replace shifter 1614 and filtering canbe done after the M¹TbIoCk. This type of implementation can be more efficient depending on the inplementation.

Next, receiver 2400 preferably includes decimatois 2406 configured to decimate the data rate of the parallel bit streams. Sampling at higher rates helps to ensure accurate recreation of the data But the higher the data rate, the larger and more complex equalizer 2408 becomes. Thus, the sampling rate, and therefore the number of samples, can be reduced by decimatois 2406 to an adequate level that allows for a smaller and less costly equalizer2408.

Equalizer 2408 is configured to reduce the effects of multipath in receiver 2300. Bs operation will be discussed more folly below. After equalizer 2408, the parallel data streams are sent to de-scrambler, decoder, and de-interleaver 2410, which perform the opposite operations of scrambler, encoder, and interleaver 1506 so as to reproduce the original data generated in the transmitter. The parallel data streams are then sent to parallel to serial converter 2412, which generates a single serial data stream fiom the parallel data streams.

Equalizer 2408 uses the multipath estimates provided by correlator 2306 to equalize the effects of murtipath in receiver 2300. Si one embodiment, equalizer 2408 comprises Single-In Single-Out (SBO) equalizers operating on each parallel data stream in demodulator 2400. Si this case, each SISO equalizer comprising equalizer 2408 receives a single input and generates a single equalized output Alternatively, each equalizer can be a MuWple-Si Multiple-Out (MMO) or a Multiple-Si Single-Out (MBO) equalizer. Multiple inputs can be required, for example, when a fiequency encoder or rate controller, such as frequency encoder 1900, is included in the transmitter. Because fiequency encoder 1900 encodes information from more than one parallel data stream onto each sub-channel, each equalizers ∞mprising equalizer 2408 need to equalize more than one subchannel. Thus, for example, if a parallel data stream in demodulator 2400 comprises d(l) + d(8), then equalizer 2408 will need to equalize both d(l) and d(8) together. Equalizer 2408 can then generate a single output corresponding to d(l) or d(8) (MISO) or it can generate both d(l) and d(8) (MMO).

Equalizer 2408 can also be a time domain equalizer (IDE) or a fiequency domain eqiializer (TOE) depending on the embodiment Generally, equalizer 2408 is aTDEifihemodulatorinthetransrrite^ data streams, andaEDEifthemodulatorperfomisroMButequaSzer2408 canbeanEDEevenifTDMisusedinthe tiHismitter. Therefore, the preferred equalizer type should be taken into consideration when deciding what type ofblock terminationtouse in the transmitter. Because ofpower requirements, it is ofienprefei^ and TDM oirthe reverse Knkin awireless αmmunication system.

As with transmitter 1500, Ihe various components comprishgdemcκϊulator2400 are preferably p^ so that a single device can operate in a plurality of different systems and still maintain superior performance, which is a primary advantage of the systems and methods described herein. Accoidingly, the above discussion provides systems and methods for implementing a channel access protocol that allows the transmitter and receiver hardware to be reprogrammed slightly depending on the communication system

Thus, when a device moves fiom one system to another, it preferably reconfigures the hardware, i.e. transmitter and receiver, as required and switches to a protocol stack corresponding to the new system. An important part of reconfiguring the receiver is reconfiguring, or programming, the equalizer because multipath is a main problem for each type of system The multipath, however, varies depending on the type of S)StBm, which previously has meant that a different equalizer is required for different types of communication systems. The channel access protocol described in the preceding sections, however, allows for equalizers to be used that need only be reconfigured slightly for operation in various systems, a Sample Equalizer Embodiment

Figure 25 illustrates an example embodiment of a receiver 2500 illustrating one way to configure equalizers 2506 in accordance with the systems andmethods described herein Before discussing the configuration of receiver 2500, it should be noted that one way to configure equalizers 2506 is to simply include one equalizer per channel (for the systems and methods described herein, a channel is the equivalent of a sub-channel as desαibed above). A correlator, such as correlator 2306 (figure 23), can ftren provide equalizers 2506 with an estimate of the number, amplitude, and phase of any multipaths present, up to some maximum number. This is also known as the Channel Impulse Response (CIR). The maximum number of multipaths is determined based on design criteria for a particular implementation. The more multipaths included in the CIR the more path diversity the receiver has and the more robust communication in Hie system will be. Path diversity is discussed a little more fully below.

If there is one equalizer 2506 per channel, Ihe CIR is preferably provided directly to equalizers 2506 from the correlator (not shown). If such a correlator configuration is used, then equalizers 2506 can be run at a slow rate, but the overall equalization process is relatively fast For systems with a relatively small number of channels, such a configuration is therefore Referable. The problem, however, is that tfieie is large variances in the number of channels used in different types of communication systems. For example, an outdoor system can have has many as 256 channels. This would require 256 equalizers 2506, which wouldmake the receiver design too complex and costly. Thus, for systems with alot of channels, the configuration illustrated in figure 25 is preferable. Bi receiver 2500, multiple channels share each equalizer 2506. For example, each equalizer can be shared by 4 channels, e.g., Chl-Ch4, Ch5-Ch8, etc., as illustrated in figure 25. In which case, receiver 2500 preferably comprises a memory 2502 configured to stare information arriving on each channel.

Memory 2502 is preferably divided into sub-sections 2504, which are each αMguredto store information for a particular subset of channels. Information for each channel in each subset is Ihen alternately sent to the appropriate equalizer 2506, which equalizes the information based on tfie QR. provided for ihatcharmeL M Ihis case, each equalizer must run much faster than it would if ttiere was simply one equalizer per channel. For example, equalizers 2506 would need to run 4 or more times as fast in order to effectively equalize 4 channels as opposed to 1. In addition, exframemory 2502 is required to buffer Ihe channel information. But overall, Hie complexity of receiver 2500 is reduced, because there are fewer equalizers. This should also lower the overall cost to implement receiver 2500.

Preferably, memory 2502 and the number of channels that are sent to a particular equalizer is programmable. Ih tins way, receiver 2500 can be reconfigured for Ihe most optimum operation for a given system. Thus, if receiver 2500 were moved from an outdoor system to an indoor system wMi fewer channels, then receiver 2500 can preferably be reconfigured so that there are fewer, even as tew as 1, channel per equalizer. The rate at which equalizers 2506 are run is also preferably programmable such that equalizers 2506 can be run at 1he optimum rate for the nur^ equalized

In addition, if each equalizer 2506 is equalizing multiple channels, then the CIR for those multiple paths must alternately be provided to each equalizer 2506. Preferably, therefore, amemory (not shown) is also included to bufier the CIR infønnation for each channel. The appropriate QR information is Ihenserώto each eq

(not shown) when the corresponding channel information is being equalized. The CIR memory (not shown) is also preferably programmable to ensm^φiimumo^

Returning to the issue of path diversity, the number of paths used by equalizers 2506 must account for the delay spread d_s in the system. For example, if the system is an outdcx>r system operating in 1he 5 GigaHeriz (GBz) range, the communication channel can comprise abandwidth of 125 Mega Hertz (MHz), e.g., the channel can extend fiom 5.725 GHzto 5.85GHz. If the channel is divided into 512 sub-channels witharoll-offractorr of .125, then each subchannel will have a bandwidth of approximately 215 kiloherlz (KHz), which provides approximately a 4.6 microsecond symbol duration. Since the worst case delay spread d_s is 20 microseconds, the number ofpaflis used by equalizers 2504 can be set to arnaximurn of 5. Thus, there would be a first pathPl at zero microseconds, a second pathP2 at 4.6 microseconds, a third pathP3 at 92 microseconds, a fourthpathP4 at 13.8 microseconds, andfifihpathP5 at 18.4 microseconds, whichis close to the delay spread c& Ih another embodiment, a sixth path can be included so as to completely cover the delay spread 4; however, 20 microseconds is ftie worst case. Ih tact, a delay spread d_s of 3 microseconds is a more typical value. In most instances, therefore, the delay spread d_s will actually be shorter and an extra path is not needed. Alternatively, fewer sub-channels canbe used, thus providing alarger symbol duration, instead of using an extrapath. But again, this would typicallynot be needed.

As explained above, equalizers 2506 are preferably configurable so that they can be reconfigured for various communication systems. Thus, for example, the number of paths used must be sufficient regardless of the type of communication system. But this is also dependent on the number of sub-channels used. If, for example, receiver 2500 went fiom operating in the above described outdoor systerntoan indoor systerO, where the dekyspread^is on 1he order of 1 microsecond, then receiver 2500 can preferably be reconfigured for 32 sub-channels and 5 paths. Assuming the same overall bandwidth of 125 MHz, the bandwidth of each sub-channel is approximately 4 MHz and the symbol duration is approximately 250 nanoseconds.

Therefore, there will be a first path Pl at zero microseconds and subsequent paths P2 to P5 at 250ns, 500ns, 750ns, and 1 microsecond, respectively. Thus, the delay spread ds should be covered for the indoor environment Again, the 1 microsecond delay spread ds is worst case so the 1 microsecond delay spread d_s provided in the above example will often be more than is actually required. This is preferable, however, for indoor systems, because it can allow operation to extend outside of the inside environment, e.g., just outside the building in which the inside environment operates. For campus style environments, where auseris likely to be tøvelingbetweenbuildings, this canbe advantageous. 7. Sample Bmbodiment of a Wireless Communicalion device

Figure 26 illustrates an example embodiment of a wireless rømmunication device in accordance with Hie systems and methods described herein. Device 2600 is, for example, a portable communication device configured for operation in aplurality of indoor and outdoor communication systems. Thus, device 2600 comprises an antenna 2602 for transmitting and receiving wireless communication signals over a wireless communication channel 2618. Duplexor 2604, or switch, can be included so that transmitter 2606 and receiver 2608 can bofli use antenna 2602, while being isolated from each other. Duplexors, or switches used for this purpose, are well known and will not be explainedhereia

Transmitter 2606 is a configurable transmitter configured to implement the channel access protocol described above. Thus, transmitter 2606 is capable of transmitting and encoding a wideband communication signal comprising a plurality of sub-channels. Moreover, transmitter 2606 is configured such that the various sub-cornponents that comprise transmitter 2606 can be reconfigured, or programmed, as described in section 5. Similarly, receiver 2608 is configured to implement the channel access protocol described above and is, therefore, also configured such that the various sub¬ components comprising receiver 2608 canbe reconfigured, or reprogrammed, as described in section 6.

Transmitter 2606 andreceiver 2608 are interfaced withprocessor 2610, which can comprise various processing, controller, and/or Digital Signal Processing (DSP) circuits. Processor 2610 controls the operation of device 2600 including encoding signals to be transmitted by transmitter 2606 and decoding signals received byreceiver 2608. Device 2610 can also include memory 2612, which can be caifigured to store operating instructions, e.g., firmware/software, usedbyprocessor2610to control the operation of device 2600.

Processor 2610 is also preferably configured to reprogram transmitter 2606 and receiver 2608 via control interfaces 2614 and 2616, respectively, as required by the wireless ∞mmunication system in which device 2600 is operating. Thus, for example, device 2600 can be configured to periodically ascertain the availability is a preferred communication system. If the system is detected, then processor 2610 can be configured to load the corresponding operating instruction from memory 2612 and reconfigure transmitter 2606 and receiver 2608 for operation in the preferred system.

For example, it may preferable for device 2600 to switch to an indoor wireless LAN if it is available. So device 2600 may be operating in a wireless WAN where no wireless LAN is available, while periodically searching for the availability of an appropriate wireless LAN. Once the wireless LAN is detected, processor 2610 will load the operating instructions, e.g., the appropriate protocol stack, for the wireless LAN environment and will reprogram transmitter 2606 and receiver 2608 accordingly. In this manner, device 2600 can move from one type of communication system to another, while maintaining superiorperformance.

It should be noted that abase station configured in accordance with the systems and methods herein will operate in a similar manner as device 2600; however, because the base station does notmove from one type of systemto another, there is generally no need to configure processor 2610 to reconfigure transmitter 2606 and recdver 2608 for cperation in accordance with the operating instruction for a different type of system. But processor 2610 can still be configured to reconfigure, or reprogram the sub-components of transmitter 2606 and/or receiver 2608 as required by the operating conditions within Hie system as reported by communication devices in cc∞rnunication with the base station. Moreover, such a base station can be configured in accordance wMi Ihe systems and methods described herein to implement more than one mode of operation. Ih which case, controller 2610 canbe configuredto reprogram transmitter 2606 and receiver 2608 to implement the appropriate mode of operation. 8. Bandwidthrecovery

As described above in relation to figures 11-14, when a device, such as device 1118 is near the edge of a communication cell 1106, it may experience interference fiom base station 1112 of an adjacent communication cell 1104. Inihis case, device 1118 will report alow SIR tobase station 1114, which will cause base station 1114toreduce the number of sub-channels assigned to device 1118. As explained in relation to figures 12 and 13, this reduction can comprise base station 1114 assigning only even sub-channels to device 1118. Preferably, base station 1112 is correspondingly assigning only odd sub-channels to device 1116.

In this manner, base station 1112 and 1114 perform complementary reductions in the channels assigned to devices 1116 and 1118 in order to prevent interference and improve performance of devices 1116 and 1118. The reduction in assigned channels reduces the overall bandwidth available to devices 1116 and 1118. But as described above, a system implementing such a complementary reduction of sub-channels will still maintain a higher bandwidth than conventional systems. Still, it is preferable to recover Hie unused sub-channels, or unused bandwidth, created by the reduction of sub-channels in responseto alowreportedSIR

One method for recovering the unused bandwidth is illustrated in the flow chart of figure 27. First, in step 2702, base station 1114 receives SIR reports for different groups of sub-channels fiom device 1118 as described above. If the group SIR reports are good, then base station 1114 can assign all sub-channels to device 1118 in step 2704. Iζhowever, some of the group SIR reports received in step 2702 are poor, then base station 1114 can reduce Ihe number of sub-channels assigned to device 1118, e.g., by assigning only even sub-channels, in step 2706. At the same time, base station 1112 is preferably performing a complementary reduction in the sub-channels assigned to device 1116, e.g, by assigning only odd sub-channels.

At this point, each base station has unused bandwidth with respect to devices 1116 and 1118. Torecoverihis bandwidth, base station 1114 can, in step 2708, assign Ihe unused odd subn;bannebto device lllδhadjacent cell 1104. It should be noted that even though cells 1102, 1104, and 1106 are illustrated as geometrically shaped, non-overlapping coverage areas, the actual coverage areas do not resemble ύiese shapes. The shapes are essentially fictions used to plan and describe a wireless communication s)dem 1100. Therefore, base station 1114 can in feet communicate with device 1116, even Ihoughit is in adjacent cell 1104.

Once base station 1114 has assigned the odd sub-channels to device 1116, in step 2708, base station 1112 and 1114 communicate with device 1116 simultaneously over the odd sub-channels in step 2710. Preferably, base station 1112 also assigns the unused even sub-channels to device 1118 in order to re∞verihe unused bandwidth in cell 1104 as welL

In essence, spatial diversity is achieved by having both base station 1114 and 1112 communicate with device 1116 (and 1118) over the same sub-channels. Spatial diversity occurs when the same message is transmitted simultaneously over statistically independent communication paths to the same receiver. The independence of the two paths improves the overall immunity of the system to lading. This is because the two paths will experience different lading effects. Therefore, if the receiver cannot receive the signal over one pafli due to fading, Hien it will probably still be able to receive the signal over theottierpath, because Hie fading lhat effected the first path wffl not effedihe second Asa result, spatial diversity improves overall system performance by improving the Bit Error Rate (BER) in the receiver, which effectively increases fee deliverable data rate to the receiver, i.e., increase the bandwidth.

For effective spatial diversity, base stations 1112 and 1114 ideally transmit the same information at the same time over the same sub-channels. As mentioned above, each base station in system 1100 is configured to transmit simultaneously, i.e., system 1100 is a TDM system with synchronized base stations. Base stations 1112 and 1114 also assigned Hie same sub-channels to device 1116 in step 2708. Therefore, all that is left is to ensure tfiat base stations 1112 and 1114 send the same information. Accordingly, the information communicated to device 1116 by base stations 1112 and 1114 is preferably coordinated so that the same information is transmitted at the same time. The mechanism for enabling this coordination is discussed more fully below. Such coordination, however, also allows encoding that can provide furfliαperfbrmance enhancements within system 1100 and allow a greater percentage of the unused bandwidth to be recovered.

One example coordinated encoding scheme that can be implemented between base stations 1112 and 1114 wife, respect to communications with device 1116 is Space-Time-Coding (STQ diversity. STC is illustrated by system 2800 in figure 28. In system 2800, transmitter 2802 transmits a message over channel 2808 to receiver 2806. Simultaneously, transmitter 2804 transmits amessage over channel 2810 to receiver 2806. Because channels 2808 and 2810 are independent, system 2800 will have spatial diversity with respect to communications from transmitters 2802 and 2804 to receiver 2806. M addition, towever, the dafotransm to also provide time diversity. The following equations illustrate the process of encoding and decoding data in a STC system, such as system 2800.

First, channel 2808 canbe denoted h_n and channel 2810 canbe denotedg;,, where: h_n =a_he/^Qh ; and (1)

Zn = ayP* (2)

Second, we can look at two blocks of data 2812a and 2812b to be transmitted by transmitter 2802 as illustrated in figure 28. Block 2812a comprises N-syrnbols denoted as aa a_\, az ..., am, or a(0:N-l). Block 2812b transmits N- symbols of data denoted b(0: N-I). Transmitter 2804 simultaneously transmits two block of data 2814a and 2814b. Blcck2814aisthenegativemveiseα)rijugateo±b^^ Block2814b is Hie inverse conjugate ofblock 2812a and canihereforebe desαibed as a*(N-J:0). It shouldbe noted that eachblock of data in the forgoing description will preferably comprise a cyclical prefix as described above.

Whenblocks 2812a, 2812b, 2814a, and 2814b are received mreceiver 2806, they are combined and decoded in the following manner: First, the blocks will be cαubined in the receiver to form the following blocks, after discarding the cyclical prefix:

Blockl =a(0:N-l) ®h_n-b*(N-l:0) <%,; and(3)

Block2=b(0:-N-1) Θh_n +a*(N-l:0) Qg_n. (4) Where the symbol ® represents a cyclic convolution.

Second, by taking an IFFT of the blocks, the blocks canbe described as: Blockl =A_n*H_n-B_n* _* G_n, and (5)

Bbck2=B_n*H_n-A_n*_*G_n. (6)

Where«=0toiV-i.

In equations (5) and (6) H_n and G_n will be known, or can be estimated. But to solve the two equations and determine A_n and B_n, it is preferable to turn equations (5) and (6) into two equations with two unknowns. This can be achieved using estimated signals^ and Y_n as follows:

X_n =A_n * H_n-B_n* _* G_n; and (J)

Y_n =B_n * H_n + A_n* _*G_n. (8)

To generate two equations and two unknowns, the conjugate of Y_n can be used to generate the following two equations:

X_n= A_n * H_n-B_n* _*G_n;anά (9)

Y_n* =B, *_*H, *+A_{n *}G_n*. (10)

Thus, the two unknowns are4i mάB_n* and equations (9) and (10) define amatrix relationship in terms of these two unknowns as follows:

Which canbe rewritten as:

Signals A_n and B_n can be determined using equation (12). It should be noted, that Hie process just described is not the only way to implement STC. Olher methods can also be implemented in accordance wilh the systems and methods described herein. Importantly, however, by adding time diversity, such as described in the preceding equations, to the space diversity already achieved by using base stations 1112 and 1114 to communicate with device 1116 simultaneously, the BER canbe reduced even further to recovσ evenmore bandwidth.

An example transmitter 2900 configured to communicate using STC in accordance with the systems and methods described herein is illustrated in figure 29. Transmitter 2900 includes a block storage device 2902, a seriako- parallel converter 2904, encoder 2906, and antenna 2908. Block storage device 2902 is included in transmitter 2900 becausea 1 blcckdekyisnec^ssarytoirnplementihecxxingillustratedinfi This is because transmitter 2804 first transmits -b_n*(n -N-l to 0). But b_n is the second block, so iftransmitter 2900 is going to transmit -δ/ first, it must store two blocks, e.g.,α,, and b_m and 1hengenerateblock28l4a and 2814b (see figure 28).

Serial-to-paraM converter 2904 generates parallel bit streams from the bits ofblocks a_n and b_n. Encoder 2906 then encodes the bit streams as required, e.g., encoder 2906 can generate -b_n* and a_n* (see blocks 2814a and 2814b in figure 28). The encoded blocks are then combined into a single transmit signal as described above and transmitted via antenna2908.

Transmitter 2900 preferably uses TDM to transmit messages to receiver 2806. An alternative transmitter 3000 embodiment that uses EDM is illustrated in figure 30. Transmitter 3000 also includes block storage device 3002, a serial- to-parallel converter 3004, encoder 3006, and antenna 3008, which are configured to perfomi in Hie same manner as the corresponding components in transmitter 2900. But in addition, transmitter 3000 includes IEFTs 3010 to take the IFFT of the blocks generated by encoder 2906. Thus, transmitter 3000 transmits -B_n* and A_n* as opposed to -b_n* and a_n*, whichprovides space, frequency, and time diversity.

Figure31 illustrates an alternative system 3100 that also uses EDM but that eliminates the 1 block delay associated with transmitters 2900 and 3000. In system 3100, transmitter 3102 transmits over channel 3112 to receiver 3116. TransmitterSlOoteansmitsovacharmeBlMtoreceiverSlld As with transmitters 2802 and 2804, transmitters 3102 and 3106 implement an encoding scheme designed to recover bandwidth in system 3100. Si system 3100, however, the coordinated encoding occurs at the symbol level instead of the block leveL

Thus, for example, transmitter 3102 can transmit block 3104 comprising symbols α_& aj, ct₂, and αj. Li which case, transmitter 3106 will transmit a block 3108 comprising symbols -ctj*, ao* -ύ₃*, and a_∑*. As canbe seen, this is the same encoding scheme used by transmitters 2802 and 2804, but implemented at the symbol level instead of the block level, Assuchjihereisnoneedtodekycneblockbeforeiransrnitting An IEFT of each block 3104 and 3108 can then be taken and transmitted using EDM An IFEl^'SllOoftlockS^isshownmfigureSl forpuiposesofillustration.

Channels 3112 and 3114 canbe described by H_« and G_n, respectively. Thus, in receiver 3116 the following symbols willbe formed:

(Ao_*Ho)-(A₁*_*Go)

(A_3*H₃)+(A₂*,G₃).

Ih time, each symbol Z_n (H = O to 3) occupies a slightly different time location. In fiequency, each symbol^ (n = 0to3) occupies a slightly different fiequency. Thus, each symbol A_n is teansmitted over a slightly different channel, Le., H_n(n =0to3)ovG_n(n = 0to3j,\\dnchresdtsm1hea3mbinatiom above.

As can be seen, the symbol combinations formed in flie receiver are of Ihe same form as equations (5) and (6) and, therefore, canbe solved in the same manner, but withoutihe one block delay.

In order to implement STC or Space Frequency Coding (SFQ diversity as described above, bases stations 1112 and 1114 must be able to coordinate encoding of the symbols that ate simultaneously sent to a particular device, such as device 1116 or 1118. Fortunately, base stations 1112 and 1114 ate preferably interfaced with a common network interface server. For example, in a LAN, base stations 1112 and 1114 (which would actually be service access points in the case of a LAN) are interfaced with a common network interfk« server tørømecteihe L

A^ such as aPublic Switched Telephone Network (PSTN). Similarly, in a wireless WAN, base stations 1112 and 1114 are typically interfaced with a common base station control center or mobile switching center. Thus, cooidination of flie encoding canbe enabled via the common connection with thenetwo&interface server. Bases station 1112 and 1114 can then be configured to share information through this common connection related to communications with devices at flie edge of cells 1104 and 1106. The sharing of information, in turn, allows time or fiequency diversity coding as described above.

It should be noted that oftier forms of diversity, such as polarization diversity or delay diversity, can also be combined with the spatial diversity in a ∞mmunication system designed in accordance with the systems and methods descnbed herein The goal being to combine alternative forms of diversity with the spatial diversity in order to recover larger amounts ofbandwidth It should also be noted, that flie systems and methods described can be applied regardless ofhenumber ofbase stations, devices, and communication cells involved

Briefly, delay diversity can preferably be achieved in accordance with the systems and methods described herein by cyclical shifting the transmitted blocks. For example, one tiansnώter can transmit abtø^ A₂, and A₃ in that order, while flie oftier transmitter transmits the symbols in the following order A₃, AQ, A₁, and A₂. Therefore, it can be seen that flie second transmitter transmits a cyclically shifted version of the block transmitted by flie first transmitter. Further, the shifted block can be cyclically shifted by more then one symbol of required by a particular implementation 9. Modulation Scheme Si Hie description that follows, methods for implementing multiple modulation schemes are presented. While these description are presented in the context of a system involving multiple Service Access Points (SAPs), it will be understood that tfie systems and methods described will also be applicable in other environments. Ih general, the systems and methods described are not dependent on any particular system architecture, geographic layout, or type of access device.

Figure 32 illustrates a communication system 3200 comprising four SAPs 3202, 3204, 3206, and 3208. As can be seen, the coverage areas for each SAP overlap with each other. SAPs 3202, 3204, 3206, and 3208, as well as flie communication devices configured to communicate with fiie SAPS, can be configured to use a wideband channel as described above; however, in certain embodiments, system 3200 can be configured such that multiple communication

wideband channel into smaller channels or bands. For example, in one implementation, system 3200 can be configured as a single band system, dual band system, or a four band system. Depending on the embodiment, it canbe preferable for the software and hardware comprising SAPs 3202, 3204, 3206, and 3208, as well as the communication device tot

Figure 33 A illustrates a single wideband channel 3302 as described above. Thus, the 3dB bandwidth (BW) for channel 3302 can, for example, be 1.5GHz. In figure 33B, the same channel is illustrated, but this time divided into two bands 3304 and 3306. Figure 33C illustrates tie same wideband channel divided into four bands 3308, 3310, 3312, and 3314.

Dividing wideband channel 3302 into multiple bands reduces Hie bandwidth available within the coverage area of each SAP; however, it also reduces interference from adjacent SAP coverage areas, due to the longer chip period (Tc), allows for lower speed equalizers, and provides frequency diversity. Accordingly, optimal performance canbe obtained by trading off bandwidth for some of these other advantages. If each SAP 3202, 3204, 3206, and 3208, are using the same single band channel, then they must be synchronized, i.e., assigned specific time slots to avoid interference with adjacent coverage areas.

It should be noted Hiat 1, 2 and 4 bans are illustrated for simplicity. A system 3200 configured according to the systems and methods described herein can use higher numbers of bands, such as 8 or 16; however, it should be kept in mind tot asystem configured in acrørdanrewithte systems andmethcxfc system. Thus, dividing wideband channel 3302 into too many bands canbe oouπter productive. The requirements of a particular implementation sliould drive flienumberofbandsused

Figures 34-36 are diagrams illustrating example hardware embodiments tot can be implemented to achieve multi-band modulation in accordance with, flie systems and methods described herein. Ih a transmitter configured to perform multi-bandmodulation, for example, a serial-to-parallel converter 3402 canbeusedto split a stream of data 3404 into multiple streams hamanner similar Mthespecificexampleoffigure34,datastream3404is split into four data steams 3406, 3408, 3410, and 3412. The data m each stream 3406-3412 can then be mcxiulated onto a separate band 3414-3420, respectively.

Figure 35 is a diagram illustrating how each data steam 3406-3412 can be modulated onto a separate band 3414-3420, respectively. As can be seen, each data steam 3406-3412 can be shifted in frequency by multipliers 3502- 3508 sotø1heywfflreademtepϊoperband34l4-3420. Eachshifieddatasteamcanihenbepulseshapedusingpulse shapers 3510-3516, and ύiencombinedin adder 3518.

In an alternative embodiment, data steams 3406-3412 can undergo an IEFT 3610 as illustrated in figure 36. The resulting transformed data steams 3614-3620 can Ihen be passed Ihrough a poly phase filter 3612 and selectively combined into a single output 3622. In still another alternative embodiment, polyphase filter 3612 can be replaced by a parallel to serial converter such as those described above.

Depending on the embodiment, combined signal 3622 can comprise complex data, i.e., values of ±1, 0, ±j. Thus, in such embodiments, an encoder 4004 can be hcludediraat can be αrfguredtoencxxieiheieal data orto red data

Data steams 4006 and 4012 can thenbeencoded in such a fastø∞inatα^ on these data streams is only re^ As explained in tie related applications, which are incorpoiated by reference, encoding data steams 4006 and 4012 using only 1, 0, or-1 can eliminate the need for a Digital-to-Analog Converter (DAQ in tiie transmitter, as well as a corresponding Analog-to-Digital Converter (ADQ in the receiver. Elimination of the DAC can save power, which can be significant since the high data rates contemplated can result in high power consumption. Elimination of the DAC, and ADC, can also reduce implementation costs.

Figure 41 is a diagram illustrating how data streams 4006 and 4012 can be implemented so 1hat only 1he values 1, 0, and -1 are used. As can be seen, each outout actually coiiiprises two outputs. For example, real data steam 4006 can actually comprise positive data stream 4102, and negative data stream 4104. The value 1 can be represented when positive data steam 4102 is high and negative data steam 4104 is low, the value 0 can be represented when both are low, and the value -1 can be represented when positive data steam 4102 is low and negative data stream 4104 is high, as illusttatedbyihe waveforms on the right hand sideofthe figure.

In certain embodiments, the highest data rate possible can be 750Mbs. Thus, in order to get the lull data rate, e.g, 1.5GHz some encoding may be needed. In one embodiment, Ihe full data rate is achieved by inserting two Os for everytwodatabitsasillusfratedinfigure42. Thus, combined signal 3622 can actuaUycornprisedteansfomieddatasignal 4202. TheOsshouldbeaddedaccordingtoanieknown^ For example, Ihe Os canbe inserted based on random sequence generation, but the random sequence should be known to both Ihe transmitter and receiver.

The transformed data signal 3622 cantiienbe shifted and combined such that the peak of one band corresponds with the 200 of another as illustrated by waveform 4302 in figure 43. Accorfingly, if single band operation is contemplated, then all of Bandwidth (BW) can be used, e.g., each of the four bands can be transmitted by a single SAP. If multi-band operation is contemplated, 1hen wavefomi 4302 should be converted into separable bands.

For example, the system of figure 44 can be used to generated dual bands that can be selected for use by a particular SAP. Here, data stream 4402 is split into two parallel data streams: and even parallel data stream 4406 and an oddparaMdatasteam4408. Ek:h data stream is Ihencc^medwi^

4414 and 4416. These data steams are then combined to produce data streams 4420 and 4422. By including delays 4410 and 4412 and ensuring that certain bits are always zero the outoutofftietransrdtter ran be configured suchto bands appear at the transmitter output when the data streams ate frequency sMM and combined: odd band 4426 and evenband4428.

Data streams 4414 and 4416 can be controlled in several ways to ensure zeroes at the correct bit locations, but two examples, Option 1 and Option 2, are illustrated in figure 44. The circuitry 4418 used to combine can, for example, comprise adders and subtractors configured as required to combine data streams 4414 and 4416 to produce the correct outputs 4420 and 4422. Alternatively, an IFF 1^' of order 2 can be used Manotheraltemative,alookuptablecanbeused to map the inputs of data stream 4402 to the output data streams 4420 and 4422. For example, the table of figure 45 can be used to map three input bits to output bits on data streams 4420 and 4422. As can be seen, two output bits ate generated for every three input bits inύie example of figure 45.

In figure 46, an example circuit 4602 and coding scheme 4604 for generating four selectable bands is illustrated Here, the incoming data 4614 is split into four parallel streams 4616-4622, which are combined with delayed versions of each other, so as to create separable bands 4606-4612. Delayed data streams 4624-4630 are passed through IFFT 4632 and ultimately combined into a single output

It should be noted that each bit on input stream 4614 can actually be a symbol representing multiple bits. For example, a two bit symbol can be used to specify the value of a real and imaginary component Thus for example, the two bits canbe used to specify the following complex information:

— 1 01 =j 10=4 ii H

Again, if a single SAP is operating without interference, or overlap witl% another SAP, then the entire band 4302 can be used ffihere are two overlapping SAPs, or four overlapping SAPS, thenmultiple bands canbe used and selected by each SAP in order to avoid interference. This selection can be achieved by selecting which bits are going to be zero and how many parallel data streams are going to be used Accordingly, it is preferable that fee transmitter and receiver circuitry be programmable so that, for example, the number ofpataUel data steams canbe selecied as required

Because Hie high data rates contemplated increase complexity and power consumption, a low data rate mode canalsobeincludedinordertoeasethesebiirdeffiwhenihehighestdatara^ Figures 37 to 39 illustrate an example embodiment of a frame 3700 structure that can be used to achieve low data rate, multi-band modulation in accordance with the systems and methods described herein In figure 37, itcanbe seen that the frame structure includes a sync 3702, header 3704, and data 3706. The function of sync 3702 is described in some of the related applications, incorporated herein by reference. Briefly, however, sync 3702 can comprise a series of codes. In the embodiment of figure 37, Golay codes (G) are used. Thus, sync 3702 comprises a series of Golay codes (G). At the end of sync 3702, a certainnurnber of inverse codes, e.g., inverse Goaly codes (-G) are used to ensure that synchronization takes place.

The purpose of sync 3702 is to allow a receiver receiving frame 3700, to ensure that it can determine that it is synchrαiizedwiihthetransnitte But because the receiver is not originally synchronized, it dc>esrκ)tknowwbatpartofframe3700itisαπκnflydetecfing. Thus, forexample,1hereceiverwillnotkrøwwhensync 3702 ends and header 3704 begins. By including inverse Golay codes (-G), the receiver is able to determine when sync 3702 ends andheader 3704 begins.

This is illustrated by waveform 3708, which shows the out put of a correlate included in a receiver receiving frame 3700. As eachGoky<xιde(G)iscorrelateα^1herarrelatewiUouψutaspike. Once the receiver sees the negative spikes corresponding to the inverse Golay codes (-G), it will know 1hat it has reached the end of sync 3702. Multiple inverse codes are included in case one or more a missed, e.g. due to fast fading. Clearly, if required by a particular implementation, more or less inverse codes can be included at the end of sync 3702, although at least 2 should be included in case one is missed for some reason.

Header 3704 can be use dot provide Hie receiver with overhead information. Normally, header 3704 can comprise bits of information that are decode by the receiver. Here, however, each bit can be represented by a code. Ih one embodiment, for example, the same code as that used in sync 3702, e.g., Golay codes, can be used in header 3704. This allows the same circuitry to be used to decode header 3704 as is use dot decode sync 3702. The code used inheader 3704 does not, however, need to be of the same length as those used in sync 3702. For example, a shorter Golay code can be used in header 3704 as is used in sync 3702. The receiver drcuritry can, therefore be programmable to allow for detection of different length codes.

As can be seen in figure 39, the data can then be represented by segments separated by codes. Here, however, the codes should be extended codes, ie., included extensions on the front and/or back of each code to allow for better correlation of the data segments. In one embodiment for examples, extended Golay codes (GE) are used Ihat include a prefix and a suffix. The prefix can comprise copies of an end portion of the Golay code (G), e.g., the last 32 bits can be used to form the prefix. Similarly, the suffix can comprise a repeat of beginning bits, e.g., the first 32 bus, of the Golay code (G). Using extended version of the codes used to form sync 3702 and header 3704 allows for use of the same decoding circuitry.

The length of Ihe extended Golay codes (GE) should be selected so that it is much shorter than each data segment in order to keep overhead low. Thelenglhofe^hsegmentshouldbeselededsotø to a manageable level, since the receiver and transmitter will not be locked as explained in the related reference, which ate incαporated herein. Further, the sum ot any extensions used should be equal to the multipath to ensure adequate correlation in order to maintain syixbronization with the transmitter.

Each bit of data can then be represented using a code, e.g., a Golay code (G). These codes can be shorter, e.g., the same length as those used in the header. The effect on the data rate is to reduce the data rate significantly, which can savepower and overhead Forexample,if1he3dBbandwidfhis 1.33GHz, then 1he chip period will be:

Tc= l/BW=750ps.

In low datarate mode, assuming a 64 bit code is used: Tbit=64xTc,and Rb= 1/Ib= 1.33Gbs/64=20MBs.

WhereRb=thebitrate.

Thebitrate can be reduced even further by increasing the length of the code used, or using multiple short codes. The latter lias the advantage Ihat each code is the same length as those used in the header.

Further, each bit can actually be a symbol representing, e.g., twice the data For example, in Quadrature Phase Shift Keying (QPSK) system, where there is both I and Q data, each symbol will carry twice the data, e.g.:

GG= 1+j G-G=I-J -GG=-l +j -G-G= -H

Here Ihe datarate is actually doubled, e.g.,Rs=2xRb=40MBs.

Accordingly, high and low data rate, multi-band modulation is achievable using the systems and methods described above. While embodiments and implementations of the invention have been shown and described, it should be apparent that many more embodiments and implementations are wilhin tie scope of the invention Accordingly, Ihe invention is nottobe restricted, except in light of the claims andflieir equivalents.

Claims

CLAIMSWhat is claimed is:

1. A transmitter tor muM^andcommu^ a splitting circuit configured to receive a serial data stream and to φHt the serial data stream into aplurality of parallel data streams; apliasesfoifecommiMiratingwithte some ofthepluralityofparallel data streams; a comhiner communicating with the phase shifter, Ihe combiner configured to combine the plurality of parallel data streams into acombined data stream; and a controller configured to select which of the plurality of data streams will be used tor ∞rnmunicationby the tHismitter.

2. The transmitter of claim 1, whaeh the splitting circmt∞rr^

3. The transmitter of claim I,wherein1hephaseshifecx)raprisesapluralityofmdtipfiers, each of theplurality of multipliers configured to multiply each of the plurality of data streams with aphase shifting signal.

4. The transmitter of claim3, further comprising apluralityofpulse shapers configuredto shape the phase shifted data streams.

5. The transmitter of claim 1, wherein the splitting circuit, phase shifter, and combiner areprogrammable, and wherein the controller is configuredtoprogramihe splitting cάciήt, phase shife, and coni)mer based on howrnany bands the transmitter is using tor communication.

6. The transmitter of claim 1, further cαiprisingaplurality of delay circuits and aplurality of adders communicating with the splitting circuit, theplurality of delay circuits andtheplurality of adders configured to combine

theplurality of adders, the combining circuitry configured to combine theplurality of combined parallel data streams

7. The transmitter of claim 1, wherein the combined data stream comprises complex data, and wherein the transmitter further comprises an encxDderoonfigured to encode Ihe cornplex data into values of 1,0, and -1.

8. Thetransmitterofclaim7,whaeintheencoderiscαifiguredtoenc^ imaginary data streams.

9. The Iransmitterof claim 8, wherein each ofthereal andimaginary datastreams αmprisestwo datastreams, wherein one of the data streams is used to indicate a value of 1, and the other data streams is usedtoindicateavalue of -1.

10. Atransntifeformulti-bandcommum^ a splitting circuit configured to receive a serial data stream andtospHt the serial data stream into aplurality of parallel data streams; aphase shifter ∞mmunicating with the splitting drcuit,1hephaseshifecxMguredto shift thephase of at least some of thepkiratityofparallel data streams; acombhercommuricating with 1he phase sl^^ data steams into a combined data stream, the αsmbined data stteamccanprismg complex data^ an encoder configured to encode the complex data of the cxjmbined data stream into values of 1,0, and -1.

11. The transmitter of claim 19, wherein the enroderis∞nfigiπedtoenαxteiheccn^ imaginary data streams.

12. Thetransmitierofclaim20,whereheachof^^ wherein one ofthe data streams is used to indicate a value of 1, and the other data streams is usedtoindicateavalue of -1.

13. The transmitter of claim 10, fttrther coirprising aplurality of delay rircuits aid aplurality of adders communicating with the splitting circuit, theplurality of delay circuits andiheplurality of adders configured to combine each of the plurality of parallel data streams with a delayed version of itself, and combining circuitry coupled with the plurality of adders, the combining circuitry configured to combine ftie plurality of combined parallel data streams accoiding to how many bands the transmitter is using for communication.

14. The transmitter of claim 10, whereiniheφKtting circuit cxm^^

15. Theiransmitter of claim 10, wherein thephase shifter comprises aplurality of multipliers, eachoftheplurality of multipliers configured to multiply each offhepluralityof data streams wifhaphaseshiflingsignaltopioάiiceaplurality of phase shifted data streams.

16. The transmitter of claim 15,fiΛerαmprishgaplι^tyofpufe data streams.

17. The transnώterof claim 10, whαehte wheamfliecGnfroUerisccαifiguredtop^ bands the transmitter is using for communication